In recent years, a great interest has appeared on developing compact and low cost "intelligent antennas" for modern commercial wireless systems. Active printed antennas stand as a promising technique in this field, as they are able of combining the radiating properties of printed antennas with the signal processing capabilities of active devices.

In the search for solutions that allow reducing the occupied space, reconfigurable multifunctional active radiators represent one of the most challenging areas in the research community. The reconfigurability is achieved thanks to the capability of these antennas to switch among different performances, optimizing the use of the limited space.

The use of active antennas permits to implement frequency translating functions, as multiplying or mixing, representing an ideal solution to implement small and cheap antenna receivers to be employed in the mobile terminals. Nevertheless, it is necessary to take into account the DC power consumption associated to many of these structures which implies the requirement of on board batteries and reduces the useful antenna life.

In this paper, a novel unbiased subharmonic mixing antenna for wireless receiver applications is proposed. A printed antenna which radiates at 1850 MHz frequency band is integrated with a subharmonic mixer based on the particular nonlinear behavior of the Enhancement mode PHEMT devices. In Fig.1a, the schematic circuit of the receiver can be seen. The frequency of the local oscillator is 920 MHz and the optimal input power of 4 dBm, the IF frequency band is about 10 MHz and the insertion losses of the subharmonic mixer are 5.7 dB.

The printed antenna behaves as a low-pass filter at IF frequencies, a band-rejection filter at LO frequencies and an antenna at 1850 MHz band. As can be seen in Fig. 1a, a simple lumped-element low-pass filter has been added at IF port, obtaining a gain of 8.1 dBi at 1850 MHz. Experimental results of the normalized antenna gain and the return losses can be seen in Fig. 1b.

An important aspect to be pointed is the unbiased characteristic of the receiver, this fact along with the compactness make it portable and independent of the batteries. Also, by injecting an IF signal through a 0° hybrid transformer to the 180° hybrid outputs (see Fig. 1a) it is possible working in an active mixing mode. In this case, by switching the drain bias, the system can be operated as a transceiver.

As mobile portable devices such as laptop computers acquire multi-mode wireless connectivity from both satellite-based and terrestrial-based transmitters, there is an increasing need for antennas having polarisation agility capabilities. We present here a reconfigurable hybrid coupler circuit as a patch antenna feeder. This system can alternatively radiate orthogonal linear polarisation, or, reverse circular polarisation waves upon changing the states of the hybrid coupler.
Fusco et. al. reported [2] that a miniature -3dB hybrid can already be synthesized using lumped components as the vertical branches of a traditional coupler (Fig.1a). Assuming that the electrical length of the series horizontal lines provides a 45

phase shift and also that impedance Z1 is equal to Zc, it is then possible to determine the value of the S21 parameter (eqn. 1).

We propose here to use a MEMS devices as bi-state capacitors in order to achieve different behaviours for the coupler circuit. In the down state, MEMS capacitors are set to provide Y2=jY1. An input signal at Port-1 will then be equally divided between ports N°2 and N°3 with a quadrature phase shift. In the upper state, the resulting capacitance of the MEMS is minimum. If we consider that |Y2| is equal to jY1/20 [1], S21 is found to be -0.034dB from eqn. 1: the hybrid circuit will appear to consist of two uncoupled parallel ë/8 transmission lines and all of the power injected into Port-1 will be transmitted to Port-2.

A first prototype operating at 5.6 GHz and using lumped capacitors was designed and fabricated in microstrip technology on a 0.15mm-thick duroid substrate (år =2.22, Fig.1b). A capacitance value of 0.568 pF was calculated for the MEMS down state. Simulations and measurements with normalised 0.5 pF capacitors are shown in Figure 2a. The magnitude of the feeding signal is evenly divided between Port-2 and 3 and a quadrature phase shift was observed. The Port-1 return loss and the Port-4 to 1 isolation are better than -20dB. The simulated and measured S-parameters without any capacitors are presented in Figure 2b. Good isolation levels are obtained between Port-1, 3 and 4 while the Port-1 to 2 insertion loss is always better than -0.3dB.

In the future, the outputs of the coupler are intended to feed a patch antenna via orthogonal slots. When the MEMS capacitors are in the upper state, each slot will directly be fed by a transmission line. By choosing which port is excited (1 or 4 or both in phase), horizontal, vertical or diagonal linearly polarised waves will be achieved. With the MEMS capacitors in the down state, the two orthogonal modes of the patch will be excited with a 90° phase difference resulting in a CP radiated wave, right or left handed CP sense is then selected by simply changing which of the coupler's input is fed.

[1] J. B. Muldavin, G. M. Rebeiz, "High Isolation CPW MEMS Shunt Switches Part 1: Modeling", IEEE Trans. Microwave Theory Tech., Vol.48, pp. 1045-1052, Jun. 2000,
[2] S. B. D. O'Caireallain, V. F. Fusco, "Quasi-Lumped Element Quadrature Coupler Design", MOTL, Vol.2, No. 6. Jun. 1989,

The design of antennas for mobile terminals pose a great challenge due to the conjunction of several conflicting requirements: lightweight, small size, sturdiness, wideband and/or multi-band operation, and low cost. In addition, the high sensitivity of small antennas to the ambience in which they operate conflicts the usage expected from a mobile terminal, which typically is surrounded by a great variety of objects. Therefore, robustness against variations in environmental conditions proves a key aspect to the successful operation of such devices.

Reconfigurable compact antennas have received considerable attention over the last years. They have been used primarily to control intrinsic operational characteristics such as: frequency response, polarization, and radiation pattern by a variety of means: utilization of electronic switches such as PIN diodes, Field Effect Transistors (FET); electromechanical switches such as relays, and microelectromechanical systems (MEMS); and silicon substrates in which plasma regions with high conductivity are created by the injection of DC current. The predominant trend in literature is that of resorting to well-known structures whose operational principle is tuned via the reconfigurability mechanism to obtain the desired control.

The advances in computational electromagnetism together with the constant increase in the computational power available to the medium user have made iterative automated design practical. In particular, Genetic Algorithms have recently found widespread use in the automated design of electromagnetic structures. They are well suited to problems in which several conflicting requirements are to be met in complex search spaces, and usually provide solutions not obtainable by conventional design procedures. Moreover, they appear able to handle the ill-posedness and customary abundance of local minima in the antenna design problem.

This work presents what to our knowledge is the first attempt to design a reconfigurable radiating element that besides being tunable in a wide band, is able to deal with environmental variations. Reconfigurability is achieved via MEMS switches located at specific positions inside a radiating structure. The geometry of the structure and the position of the switches are simultaneously optimized by a Genetic Algorithm seeking the best performance with low structural complexity under several operational characteristics, representative of actual application scenarios.

Design, simulation, and fabrication of a novel floating broadband coplanar waveguide (CPW)-fed spiral on-chip antenna using silicon micromachining will be presented. The antenna is designed for unlicensed 60 GHz band, i.e., 57-64 GHz for millimeter-wave applications. To maximize the radiation efficiency of the antenna, we have used silicon bulk micromachining method. The KOH wet etching method used in this work is capable of fabrication of membrane areas as large as 4 mm by 4 mm. Simulation results of the antenna using Finite Integral Technique (FIT) will be presented. The antenna has more than 15 GHz impedance bandwidth and has directive radiation pattern at 55 and 60 GHz and its directivity is 6.359 dBi and 4.944 respectively.

(a) (b)(c)
Fig. 1: (a) SEM image of the patterned cavity created in the backside of the silicon substrate, SEM image of the fabricated floating antenna, (b) on wafer view, and (c) backside view.


Fig. 2: Optical images (step by step) to realization of floating on-chip antenna .


(a) (b) (c)
Fig. 3: Simulation model (a) on wafer view, (b) backside view, and (c) S11 parameter of the antenna .


(a) (b) (c)
Fig. 4: Radiation pattern of the antenna (a) axis on the antenna (b) Directivity at 55 GHz (c) Directivity at 60 GHz

Radio frequency micro electrical mechanical systems (RF MEMS) are an emerging technology that promises the potential of revolutionizing RF and microwave system implementation for the next generation of telecommunication applications. Typically, RF MEMS switches are classified into two types: resistive series and capacitive shunt switches. They are normally built on high-resistivity silicon wafers, gallium arsenide (GaAs) wafers, and quartz substrates using semiconductor microfabrication technology with a typical four-to six-mask level processing. In this paper, we proposed a triple band reconfigurable antenna using RF MEMS switches that can be fabricate in easy process steps.We have designed and simulated patch antenna that can be operated in three different frequency bands depending on the MEMs switch conditions. The proposed antenna can be tuned at 17.36, 23.54 and 29.87 GHz.

The schematic diagram of the reconfigurable antenna is shown in Fig. 1. It consists of three patches where the feeding configuration exists in the center patch. The patch antenna design was supported with a model built using a high frequency structure simulator based on finite elements modeling (FEM). In our proposed fabrication process, the patch antenna is fabricated by usual method and the MEMS switch will be fabricated on a dielectric layer by metal deposition separately. We then apply sandwich method to placed the MEMS switch on the patch antenna that can be bonded by benzo cyclo butene (BCB) adhesive layers, which have good electrical properties at high frequencies as well as adhesive properties for the bonding process [1].

Fig. 2 shows the response of the antenna when both switches are ON and OFF, which indicates the antenna can be operated in reconfigurable ways.

Conclusions

A reconfigurable triple-band antenna operating at 17.36, 23.54 and 29.87 GHz using MEMS switches is presented. Single resonant frequency occurred when both the MEMS switches are ON confirmed that the antenna behaves like a single patch antenna. The proposed method can be implementing in easy fabrication process steps by sandwich methods.

References

[1] Jung-Mu Kim, et. al. "Silicon MEMS probe using a simple adhesive bonding process for permittivity measurement" J. of micromechanics and microengineering, 2005.

A planar active integrated antenna array with a high power amplifier circuit was made with a creased circuit substrate for effective heat sink. The high power amplifier circuit consists of three-stage amplifiers with a couple of watts as the output power. Due to conversion loss from DC to RF power in the second and third amplifier, the heat sink was requested. A metal (aluminum) plate was attached with the creased circuit substrate and creates heat flow path within the light active integrated antenna plate. The three-stage amplifier was made in the soft circuit substrate creased and bent with a right angle. This configuration enables us to easily make a large 2D active integrated antenna array plate for high power wireless power transmission.

I. Introduction

The increasing demand for communication systems has called for the design of low-cost and small-size radio frequency and microwave transceivers. One of the design approaches is to integrate different parts of components into single element such that fewer components are to be used. To realize a high date rate of wireless transmission, it is necessary to use a wideband antenna, and to suppress the interference signals in this band one needs a band-stop filter. If the antenna package is to be installed in a portable wireless device and communication system, the package must be compact in size.

The objective of this paper is to describe a combination of antenna and L-resonator band-stop filter useful for such applications. The parameters which affect the operation of the antenna-filter device analyzed both numerically and experimentally.

II. Antenna-Filter Design

Planar monopoles with various radiator shapes such as circular, square, elliptical, pentagonal and hexagonal, have been widely discussed and used. These broadband monopoles feature wide operating bandwidths, satisfactory radiation properties, simple structure and easy of fabrication. However they are not planar structures because their ground planes are perpendicular to the radiators. As a result, they are not suitable for integration with printed circuit board. This drawback limits practical applications of these broadband monopoles.

Fig.1 shows the geometry of the proposed antenna-filter device. A rectangular monopole with a patch size,18x20mm^2, and a 50 ohms microstrip feed line are printed on the same side of the dielectric substrate (in this study, the Teflon substrate of thickness 0.76mm and the relative permittivity 3.5 was used). L (=45mm) and W (=28mm) denote the length and the width of the dielectric substrate, respectively. On the other side of the substrate, the conducting ground plane with a length of Lg (=22.5mm) only covers the section of the microstrip feed line. H (=2.5mm) is the height of the feed gap between the feed point and the ground plane. Also, the band-stop element with an open-circuited L-resonator or a short-circuited L- resonator is introduced on the opposite side of the dielectric substrate and is coupled to the ground part (or imaging part) of the main planar monopole antenna. Here, G is the distance of the coupling gap between the resonator and the ground plane. Lr and Wr denote the length and the width of the resonator, respectively. The addition of a L-resonator element makes a band-stop characteristic.
Fig.1 The geometry of the proposed wideband antenna with L-resontor band-stop filters

Some passive millimeter-wave harmonic mixers with integrated LO-IF diplexers have been designed and characterized for both 77 GHz and 120 GHz frequency bands. These modules have been employed to work as an extension for the higher millimeter-wave antenna measurements, particularly the radiation pattern measurements. Through this article, several characterized parameters have been presented, such as the conversion loss, the IF and LO return loss and the LO-IF isolation, besides the design and simulation procedures of the integrated diplexer and the RF and LO impedance matching networks of the mixer. The realized modules have been used to record the radiation pattern of two antennas operating at the same bands.

LTCC multilayer modules are appreciated for their flexibility in realising an arbitrary number of layers with easy-to-integrate circuit components like via-holes, cavities, thickfilm resistors, SMT components and chip devices. For the frequency range of Ka-Band, manufacturing tolerances become more critical with respect to the wavelength. Yet, the demand for high integration is especially increasing for this frequency range. For example, the increasing demand for mobile access to fast data services is one of the drivers for future broadband satellite systems in Ka-Band. Antennas employing Digital Beam-Forming provide fast and flexible reconfiguration required in such systems without the necessity for moving mechanical parts.

The antenna presented in this paper is a circularly polarised, 8´8 element array, consisting of four 4x4 element building blocks. The array can be used for a digital beam forming terminal (transmitting system) operating from 29.5 GHz to 30.0 GHz. The circular polarization of the antenna elements is achieved by a hybrid ring coupler. The area for the hybrid ring coupler circuit is limited to the grid defined by the cell size of one element, which is half a wavelength. The array features four calibration networks to enable an automatic array calibration of each building block consisting of 16 antennas. The area of one calibration network is limited to the size of the 4x4 element array building block, which is two wavelengths. This high density integration task is achieved by vertical integration of antennas and circuitries in an LTCC multilayer module, consisting of 11 layers.

In this paper, the simulation of the whole array and its calibration networks are presented and compared with measurements. The simulation and measurements include thorough characterisation and tolerance analysis of the complete 4x4 building block and moreover the behaviour of the complete 8x8 array, steered to different main beam angles.

For present-day wireless communication systems, the demand for increased capacity and the need of high efficiency has generated much interest in the study of antennas. The microstrip antennas exhibit a low profile and are light weight. However, major disadvantage of these antennas in practical applications is a narrow bandwidth and the much effort has gone into bandwidth enhancement techniques. The required operating bandwidths for antennas are about 7.6% for a global system for mobile communication (GSM, 890-960 MHz), 9.5% for a digital communication system (DCS, 1710-1880 MHz), 7.5% for a personal communication system (PCS, 1850-1990 MHz) and 12.2% for a universal mobile telecommunication system (UMTS, 1920-2170 MHz). The main goal of this study is to design wideband antenna array using adaptive techniques (Beamforming and Direction of Arrival estimation) for steering the main beam towards the desired signal, and creates nulls in interference directions by controlling excitation phase of each radiating element. The adaptive antenna arrays are important components of modern wireless applications, in order to improve system capacity and overall communications performance. In this paper the broadband phased-array controlled by RF phase shifters using Sequential Quadratic Programming (SQP) algorithm [1] for adaptive beamforming is designed. This phased-array is capable of forming a desired radiation pattern via tuning the appropriate phase control.

The broadband operation can be obtained by using a radiating rectangular microstrip antenna with an E-shaped patch on foam substrate [2]. The feeding is realized by coaxial probe (Fig.1). This antenna has been simulated using FDTD code and measured using a WILTRON 360 network analyzer in anechoic chamber. Fig. 2 (a) shows the simulated and measured return loss. The obtained impedance bandwidth (-10 dB) is 556 MHz or about 26.17% with respect to the designed center frequency at 2124 MHz. It is observed that this antenna is suitable for PCS and UMTS applications. The measured E-plane and the H-plane gain patterns of a single patch are shown in Fig.3.
Using the dimensions of the single patch, eight of these antennas are combined in to an array (1 x 8 patches) using a phase only Beamforming network, forming a phased-array antenna with electronically controlled excitation phase element weighting, also shown in Fig.4. The distance between each radiating element is 0.5 lamda. The optimal excitations phases of each radiating element are determined by SQP algorithm see [1]. The beam and null steering of the array antenna is controlled by vector Modulator (The IQ signals give us full phase and magnitude control). The example of radiation pattern for this phased-array is shown in Fig. 2 (b). The desired signal arriving at 40° and one interferer signal arriving at -50°.

1. M. Mouhamadou, P. Vaudon, and M. Rammal, "Smart Antenna Array Patterns Synthesis: Null Steering and Multi-User Beamforming by Phase Control" Progress In Electromagnetics Research, PIER 60, 95-106, 2006.
2. F. Yang, X.-X. Zhang, X. Ye, and Y. Rahmat-Samii, "Wide-Band E-shaped patch antennas for wireless communications" IEEE Trans Antennas Propagat 49 (2001), 1094-1100.

Uniformly spaced arrays have been consistentely employed for active phased array system.

This class of array loses its controllability toward a grating-lobe when it appears in the visible region.

As two-dimensional array optimization problem has quite a large number of degrees of freedom, it is difficult to derive a deductive method. In this study, a 2D array optimization method based on an evaluation function defined in the beamspace is presented and we introduce a fast algorithm using a potential function.

Utilizing Dirac's delta function that is positive infinite at which an antenna is placed, we have an antenna displacement function as, a_0(x) = Σ δ(x-x_n). where x_n is the position of n-th antenna.

The response b_0(h), as a function of normalized wavenumber h, of the antenna array a_0(x) is defined by a two-dimensional Fourier Transform pair:

b_0(h) ←→ a_0(x).

When the main beam is steered to a direction h', the corresponding beam function is b(h)=b_0(h-h').

Then, the visible region for b_0 is given by |h-h'| ≤ 1.

If we allow the main beam direction h' to steer within 0 ≤ theta ≤ theta_s, the envelope of the region defined by |h-h'| ≤ 1 becomes the region D: |h| ≤ 1+sin(theta_s).

In this study, we set |h| ≤ 2 since the beam can be steered from vertically to horizontally and thus 0≤theta_s≤90°. To evaluate the goodness of an array, we introduce a mean-square function in D in h-space.

Using a weight function P(h), the evaluation function is defined as,

To evaluate a given array using the evaluation function, Fourier Transform to obtain b_0(h) and a numerical integration are required.

We develop a faster optimization algorithm that does not require Fourier Transform and numerical integrations. According to Parseval's theorem, the evaluation function is also written as a function in h-space, using convolutional integration, as,

This algorithm can be performed very fast because it is not necessary to calculate an integration or Fourier Transform.

The resulting array of this algorithm has a triangular arrangement with an uniformed distance of less than 0.5 wavelenth. In order to match an electromagnetic condition that every adjacent antenna pair would be apart more than approximately 0.8 wavelength we employ another potential in addition to the potential function derived above. By applying this potential, the algorithm generates an aperiodically arranged array shown in Fig.1(right). It is shown that this array has a preferable beam pattern and is well suited for adaptive beamforming applications.

Active array antennas have been combined with cascaded Butler Matrices to produce a high gain narrow beam and highly linear broad beam active antenna system. The output signals from the first Butler Matrix (BM), which acts as a beamforming network, have high gain and narrow beam width for long distance communication. The outputs from the second BM, which acts as a mirror of the first BM, reconstructing the antenna patterns of the individual radiating elements, have high linearity and broad beam width that can be used for short distance communication.

The configuration of the system that consists of an array of patch antennas with 0.5 lamda spacing, low noise amplifiers (LNAs), Wilkinson power divider and two BMs is shown in Fig. 1. Signals amplified by identical LNAs enter the input ports of the first BM and are subdivided into equal amplitude with progressive phase variation across the output ports, for high gain, narrow beam reception. These signals are then fed into Wilkinson power dividers. One output of each Wilkinson forms a narrow beam output port while the second outputs are amplified and recombined in the second BM. When compared to a system involving direct, single ended amplification of the individual patch outputs this system offers enhanced linearity, because the outputs of the four high linearity amplifiers are power combined by the second BM. Further, the effect of the loss of the BM on the system noise figure is reduced with the introduction of LNAs before each stage of the BM.

Measured results for a passive version of the network (without the amplifiers) are shown in Fig. 2. Measured radiation patterns from Port 1 to Port 4 of the first BM are shown in Fig. 2(a). The figure clearly shows the beamforming pattern with narrow beam when a single Butler matrix is used in the system.

Fig. 2(b) also shows the radiation pattern out of the second Butler Matrix. It shows than when Butler Matrices are cascaded broad beams will appear, as the individual antenna patterns are regenerated. The peak power output from the second Butler matrix is about 10 dB lower. This is due to the cancellation of the array factor and the losses of the second Butler Matrix. The results show that when the proposed setup is implemented two types of pattern will appear. The narrow beam can be selected for long distance communication while the broad beam can be selected for shorter distances. In the active version, the second BM acts as a four-way power combining network, which will create a nominal 6 dB improvement in linearity. This setup is potentially useful for vehicle application when Inter Vehicle communication and Vehicle to Roadside communication are required simultaneously. The full paper will include full details of the measurements and supporting theory.

Fig. 1: The System Configuration

Fig. 2: Radiation Pattern of the system (a) Narrow Beam (b) Broad Beam

Some communication satellites have an antenna that consists of a fixed parabolic reflector that is fed by an array. Such antenna systems can be programmed to produce a variety of fixed beams in response to traffic demands and can be built so that each array element is fed by a its own high power amplifier.

However benefits can accrue if the amplifiers are incorporated in Multiple Power Amplifiers (MPAs). The architecture of an MPA is shown in Figure 1. Ideally all the MPA inputs are divided equally between all amplifiers and there is a one to one correspondence between the MPA feed ports and the MPA array ports.

Due to imperfections in the INET and the ONET, and differences between the amplifiers, the ideal behaviour is never realized in practice. To optimize the performance of any manufactured MPA a deliberate offset in the gain and phase shift of each amplifier has to be introduced to compensate for the imperfections introduced by the INET, the ONET and their associated cabling. This paper shows how the manufacture of an MPA has been aided by computer modelling using the Agilent ADS software and by the use of a special vector modulator to achieve the wanted gain and phase offsets.

Figure 1. MPA Architecture The paper will present the details of a model of the MPA and its relation to the antenna. A report will be given of the results obtained of a working S-band, 8-way MPA in which the systems design and passive equipment has been executed by Astrium Space and the vector modulators and amplifiers have been designed and constructed by Roke Manor Research. The paper will describe some of the results of the modeling activity and will compare them with the measured results. In addition the benefits of higher order MPAs will be demonstrated by analysis.

The work has been funded in part by the European Space Agency.

Microstrip reflectarray antennas are currently considered in satellite missions requiring low cost, small stowage volume during launch and lightweight solutions. One possible application is microsatellite-based SAR payloads [1]. In this case, it is essential to have a beam steerable antenna in which the reflectarray has phase-tunable elements. Such cells implemented with MEMS and varactor diodes are proposed in [2] and [3] respectively. In order to assess the performance of such antennas, a small scale prototype has been designed and built using the tunable unit cell described in [3]. Typically, the cells’ phase settings are determined with a theoretical model that assumes identical behaviour of all the elements of the array, and well known phase distribution of the incident illumination. Single elements tested separately in a waveguide simulator showed very similar, but slightly different characteristics. A loss budget was made on the reflectarray prototype, taking into account taper and spillover losses as well as dissipated power in materials and diodes. It was found that in the best case, there is a difference of 2.3 dB between the measured (13.3 dB) and the expected gains. It was conjectured that this loss could be attributed to a non-optimal phase profile imposed on the elements, possibly resulting from variability in the diodes, non-constant angle of incidence, fabrication errors, unexpected distortions in the incident illumination phase, etc. The object of the work to be presented in this paper is to implement an adaptive gain optimization process and determine to what level the unexpected losses mentioned above can be compensated by tuning all the scattering cells.

The microstrip reflectarray antenna is made of 30 elements placed in a 5x6 array. Each of the elements consists of an aperture-coupled microstrip patch and two varactor diodes. Different voltages are applied to those elements to ensure a maximum gain. The gain optimization algorithm presented in [4] was implemented in Matlab script. This script is driving a D/A card (Analog Device AD5532-1) providing 30 control voltages. The gain is measured at each state of the voltage vector generated by the algorithm using the sequencer capability of the antenna range controller (MI Technologies MI-3000). The final paper will describe the various optimization strategies and the resulting improvement on the achieved maximum gain.

References:
[1] H. Legay, B. Salome, E. Girard, S. Ramongassie, J. Encinar and G. Toso, "Reflectarray Antennas for SAR Missions," Proc. ANTEM 2005, pp. 472-473, St-Malo, France, June 2005.
[2] S. Avrillon, L. Mercier, A. Pothier and P. Blondy, "Reflectarray with Integrated Band Reject Filter for MEMS Based Beamforming Applications," Proc. ANTEM 2005, pp. 474-475, St-Malo, France, June 2005.
[3] M. Riel and J.-J. Laurin, "Design of a C-Band Reflectarray Element with Full Phase Tuning Range using Varactor Diodes, Proc. AP-S URSI Symp., vol. 3a, pp. 622-625, Washington DC, July 2005.
[4] T.A. Denidni and G.Y. Delisle, "A Nonlinear Algorithm for Output Power Maximization of an Indoor Adaptive Phased Array," IEEE Trans. Electromag. Compat., vol. 37, pp. 201-209, May 1995.

In this paper we introduce a new methodology to design Beam-Forming Networks - BFN for Phased Arrays systems, in which, to scan the beam, is not longer necessary to use a phase shifter for each radiating element conforming the array. The new methodology is based in the Coherently Periodic Radiating Structures (CORPS) Principles. With the use of this concepts, we can reduce the electronic necessary to scan the beam, e.g., a 3 by 3 Antenna array can be scanned using only 3 phase shifters instead 9. A prototype of this kind of networks was designed and its measurements are presented here. Using the same idea, bigger systems can be built too.

Introduction

The recently proposed CORPS are applied in this opportunity to BFN for feeding antenna arrays. The system proposed can reduce by a factor of three, the phase shifters necessary to make a hemispherical scanning of the main beam. A BFN designed based on CORPS principles, as can be seen in fig. 1, has three main characteristics: First, the network is a periodic structure, so, it can be analyzed using unit cells. Second, the network has defined a horizontal filter that prohibits the energy propagation beyond the unit cell in this plane. And third, all the energy that is present throughout the BFN are recombined in a coherent fashion, so, in the upper layers there are no information lost. To achieve these characteristics, each BFN, feeding three antennas is made based on Gysel Cells. The prototype, showing in fig. 2, consists of 3 BFN that feed each one to a linear array of three antennas, and jointly, they built a 3 by 3 antenna array. In the same way, these three BFN are interconnected to each other by 2 identical BFNs, thus we have only four inputs to the complete BFN of the array.

In traditional systems, the beam scanning is made applying a phase shift directly behind the antenna fed, these means that for scanning a 3 by 3 antenna array is necessary to use 9 phase shifters and all the electronic associated to these equipment. Our proposal reduces the use of phase shifters in a factor of three. Of the 4 input ports of our BFN, one is used as reference and, each one of the other are associated to a phase shifter. Thus, the phase and amplitude of each radiator will be dependent of the amplitude and phase shift given at input ports. As can be seen in the fig. 3, each BFN, independently of the layers that could conform it, makes a linear representation at output port of the phase shift applied to input ports, and the amplitude at output ports presents a Gaussian shape making possible obtaining scan angles of about 20 deg. for each 1x3 linear array.

Conclusion

At this point, its easy to extrapolate the results obtained here and conclude that it is possible to control bigger systems, i.e., an N by N antenna array can be scanned by only use 3 phase shifters using this design methodology.

Fig. 1. Prototype of CORPS based BFN fabricated on Glass Fiber. Working Freq. 2.75 GHz

Fig. 2. Schematic representation of a 3x3 antenna array feeding by CORPS based BFN

Fig. 3. Radiation Pattern measured for a CORPS based BFN. A scanning of about 20 deg. is obtained.

This paper deals with the design of a spotforming Ultra Wideband system using smart antenna technology. As opposed to beamforming, we propose a method to form a 'spot' at an intended location. Smart antenna systems can be used to selectively form beams and nulls at intended directions. However, if an interferer lies within the beam of a desired recipient, the interference cannot be mitigated. The advantage of forming a spot is that it reduces interference from interferers closer to the receiver despite of it being within the direction of the beam. The problem also includes the designing and selection of best suited antenna for this application. Beamforming is a well-known signal processing technique used to control the directionality of the reception or transmission of a signal on a transducer array. It narrows down the direction of perception of the signal and hence reduces the transmitted power. A smart antenna is an array of antenna elements connected to a digital signal processor. Such a configuration dramatically enhances the capacity of a wireless link through a combination of diversity gain, array gain, and interference suppression. Increased capacity translates to higher data rates for a given number of users or more users for a given data rate per user. Smart antenna transmitters can encode independent streams of data onto different paths or linear combinations of paths. Smart antenna systems can be mainly classified in two categories: (i) Switched Beam and (ii) Adaptive arrays. The process of spotforming relates to creating a spot around the mobile user. This requires an increased number of antennas for transmitting and receiving the signal. It gives an exceptionally improved signal to noise ratio as it is clearing off inline interfering signal. For creating a spot we integrate the concepts of smart antennas and UWB systems. As defined by FCC, for UWB systems, the fractional bandwidth should be greater than 0.25, i.e., (fh-fl)/fc > 25%. An alternate definition is that band width should be greater than 500MHz. Upper and lower bounds are defined where power fall 10 dB from its peak value . Numerical simulation results corroborate the concept of spotforming. Spots of different sizes have been created using different number of antennas in the array and different configurations. We have also used different types of UWB antennas to customize the shape of the spot. Using this techniques, holes can also be selectively created to mitigate interference.

Adaptive beamsteering and beamforming techniques are being introduced in modern wireless systems, as a way to reduce interference while improving link gain and system capacity. Requiring only simple analog circuits, retrodirective arrays are standing as a promising alternative when compared to other complex digital-based processing topologies.

Retrodirectivity may be assured if each array element responds with a phase conjugated version of the arriving signal. In Pon type of solutions, this is accomplished through its mixing with a local oscillator (LO) at twice the interrogation frequency [1].
In this abstract, a dual-gate resistive mixer [2] is integrated with a slot coupled square patch to be used as a retrodirective array element. The first gate DC voltage is set to pinch-off for maximum variation of the device output conductance, while V_DS is fixed to 0V. The signal arriving at drain terminal beats with the time-varying conductance to produce the phase-reversal component. Controlling the second gate voltage with a pulsed data signal, the so generated response may carry information back to the interrogator position in an Amplitude Shift Keying (ASK) modulation format.

A simplified diagram of the phase conjugating antenna test set-up is presented in Fig. 1. Details of the printed patch and the mixer schematic may be appreciated. Back at the interrogator position, an Agilent 89600 Vector Signal Analyzer allows measuring the retransmitted signal. In Fig. 2, the amplitude component of the response envelope may be observed. ASK modulation at the original 1MHz pulse rate is perfectly validated.

References

[1] C. Y. Pon, "Retrodirective array using the heterodyne technique,"IEEE Trans. Antennas and Propagat., Vol. AP12, pp. 176-180, Mar. 1964.
[2] S. A. Maas, The RF and Microwave Circuit Design Cookbook, Norwood: Artech House, 1998.

Abstract

A technique has been reported in the past for generating pseudo-metallic plasma inductive grid arrays within a semiconductor wafer [1]. A model that can present the plasma generation procedure is developed here. Through this model, plasma and temperature distributions in the silicon wafer can be calculated as a function of time. Representative results of an investigation into the transmission properties of a grid array with respect to the outcome of the plasma generation procedure are presented. The simulations are based on a full wave analysis of transmission/reflection plane wave performance of single layer planar FSSs[2].

Illumination of a semiconductor results in a temperature increase of the semiconductor and, depending on the frequency of the illuminating beam, the generation of an electron-hole plasma in the bulk of the semiconductor [3,4]. The equations governing these procedures are written as:

Equation (1) is the continuity equation for the plasma density. In this equation the rate of change of plasma density, set equal to the generation minus the recombination rate, is written considering Auger process [5] to be the dominant recombination mechanism with Auger lifetime depending upon carrier concentration according to: [6]. Equation (2) describes the temperature increase of the semiconductor due to the excess of photon energy over the semiconductor band gap , the free carrier energy absorption and the release of energy following recombination. Equation (3) describes the attenuation of the beam intensity , used for the excitation of the electron-hole pairs, due to the lattice and free carrier absorption.

A representative result of the plasma generation response with time is shown in Fig. 1. The transmission coefficient dependance on time is shown in Fig. 1, as well. The exploitation of the high concentration for a short period of time is investigated. This paper aims to clarify the dynamic characteristics, as compared to a convnentional FSS, of a photo-illuminated silicon wafer.

Figure 1 Plasma Concentration response with time for a pulsed light source

Figure 2 Transmission Coefficient Response in Time for different frequencies

References:

[1] D. S. Lockyer, J. C. Vardaxoglou, M. J. Kearney: "Transmission Through Optically Generated Inductive Grid Arrays", IEEE Trans. Microwave Theory Tech. , vol. 47, pp. 1391-1397, Jul. 1999.
[2] J. C. Vardaxoglou, Frequency Selective Surfaces: Analysis and Design. New York: Wiley 1997, ch. 2.
[3] M. G. Grimaldi, P. Baeri and E. Rimini (1984): "Laser Induced Free-Carrier Absorption in Si Single Crystal", Appl. Phys. A33, 107-111.
[4] H. M. Van Driel (1987): "Kinetics of High Density Plasmas Generated in Si by 1.06 and 0.53-ìm Picosecond Laser Pulses", Phys. Rev. B35, (15), 8166-8176.
[5] A. R. Beattie and P. T. Landsberg (1959): "Auger Effect in Semiconductors", Proc. Roy. Soc. London A249, (1256), 16-29.
[6] J. S. Blakemore (1987): Semiconductor Statistics, Dover

MIMO systems (Multiple Input Multiple Output) are gaining more popularity as the next generation wireless systems are being developed. A system containing multiple antennas at receiver and/or transmitter increases the cost, complexity, and space needed. For a given limit on the number of antenna elements, reconfigurable antennas are good candidates for increasing the performance of the system. We investigate planar antenna structures together with MEMS switches to design and build reconfigurable antenna elements that could be placed in handhelds following the MIMO paradigm. One of the key issues is the analysis and design of the antenna and how it may incorporate the integrated MEMS switch. The focus is set on the implementation of the switches on the antenna surface. With the MEMS switch integrated in the antenna structure, a more flexible design procedure is achieved. Figure 1 and 2 show two examples of reconfigurable antennas. In Figure 1, a meander slot antenna is presented after MEMS switches have been mounted. The antenna will switch between four different frequencies due to the two MEMS switches placed on the asymmetrical meander slot. This design could also be extended to include more than two switches. The possibility to cover a range of frequency bands continously then occurs. In Figure 2, a polarization reconfigurable PIFA is presented. This antenna will switch between two different polarizations when the switches connect the top patch layer to ground. Polarization diversity is then achieved with the implication of possible increase in capacity and throughput for the wireless channel. Currently, more switches are being manufactured with the aim to build more reconfigurable MEMS antennas based on slightly different structures. Simulations and measurements are presented as well as a discussion on the applicability of our antennas.

S-band multimedia broadcasting services to mobiles (S-band Digital Mobile Broadcasting, S-DMB), supplied by geostationary satellite systems via multiple shaped beams antennas for different linguistic geographical regions, have been recently proposed. Large effective aperture antennas are required to provide high data rates and quality of service to low-cost and small-size mobile terminals, with a possible implementation by means of large size reflectors fed by complex arrays (Array Fed Reflector - AFR). High RF power handling capability along with flexibility in its allocation is required as well, leading to a highly challenging payload system to be coped with.

The paper will present a promising architectural solution and optimization results of a flexible payload for S-band mobile broadcasting to multiple shaped-beams. The outcome of the design and optimization activity shows a high degree of coverage reconfigurability as well as an almost full freedom in power-to-beam allocation of the proposed solution.

The high power section, providing RF power generation and distribution to feeds with appropriate amplitude and phase to realize different shaped coverages, constitutes a key element of the proposed payload. Among possible candidate configurations characterized by different amplitude and phase beamforming constraints, the proposed architecture is based on a high-power Butler-like Hybrid-Matrix Beam-Forming Network appropriately connecting a stack of HPAs to the feeds.

The solution consists of a generalization of the Multi-Matrix canonical scheme, i.e. a stack of appropriately connected multi-port amplifiers (cascade of Input Network - INET, Amplifiers, Output Network - ONET) allowing to generate a regular lattice of spot-like beams. Since Multi-Matrix canonical scheme may not be completely suitable for contoured linguistic coverages, amplitude and/or phase degrees of freedoms at low-power INET level are introduced to better fit geographical boundaries. Two different optimization strategies can be actually selected, with different impact on radiation performance vs. power efficiency: 1) Amplitude & Phase, implemented by means of fully interconnected INET with complete control of amplitudes and phases, which leads to the best radiation performance against a non-optimum power efficiency due to the fact amplifiers usually work at different operating points. This efficiency degradation can be however traded-off for example making use of flexible amplifiers, whose flexibility range is used to constrain amplitude ripple during optimization. 2) Phase-Only, implemented by means of fully interconnected INET with control of only the phase variables supplied along with a uniform amplitude distribution, whose reduced radiation performance with respect to A&P strategy is compensated by an optimum power efficiency due to the fact that all the amplifiers work at the same operating point on a power pooling base.

Then, the purpose of the high-power Butler-like hybrid matrices after the amplifiers is a partial reshuffling of the amplitude and/or phase degrees of freedom from the hybrids input to the hybrids output, i.e. at feed level, in order to supply the proper contoured beam composition. In this respect the proposed architecture allows the optimisation of radiation performance in terms of both guaranteed minimum gain over the coverage and beam to beam isolation (in case of frequency reuse), the latter being achieved with limited impact on gain degradation.

The purpose of this investigation is to identify a set of active antenna architectures that can provide in-orbit reconfigurable contoured beams for telecommunication applications in the Ku-band of the frequency spectrum. The author analyzes in detail single and dual reflector antennas illuminated by an array of feed horns. In each case, design methods are derived with emphasis on minimizing the number of feed elements necessary to obtain a reduced size variable beam forming network. A compact multiple beam antenna is presented, based on mission requirements of a geosynchronous satellite deployed over different geographical regions, taking into account constraints due to adjacent satellite interference. The author discuss the main technical trade-offs amongst antenna architectures with separate transmit and receive functions, combined transmit and receive functions and a direct radiating arrays. The analyses are compared with that of shaped reflector antennas.

For the design of active microstrip phased antenna arrays, low cost unit cells are required, enabling the control of the phase distribution on the array without additional phase shifting circuitry. In this work, a novel variable phase-shifter is presented for its use in active microstrip phased array antennas, based on an injection-locked third harmonic self-oscillating mixer. The novel circuit design provides a phase-shift range up to 540°, which assures that the circuit can be used in a range of at least 360° without the appearance of noisy-precursors [1]. The circuit also provides a first down-conversion of the input signal to intermediate frequency. The third harmonic self-oscillating mixer is optimized for maximum conversion gain, through the use of new nonlinear design techniques [2], which are based on the use of a novel multi-harmonic load at the input of the circuit (Fig. 1). When used as a variable phase-shifter, the third harmonic self-oscillating mixer is injection-locked with a synchronization generator at the self-oscillating frequency (Fig. 1). The synchronization ranges versus the circuit parameters are obtained through the harmonic balance simulation of the synchronization loci (Fig. 2), and the corresponding phase-shifts are calculated (Fig. 3). For this analysis, use is made of an auxiliary generator (AG) [3], as presented in Fig. 1. The stability of the synchronized solutions of the circuit is analyzed through envelope transient simulation [4] (Fig. 4). An 11.3-1.5 GHz phase-shifter/down-converter with a 3.25 GHz free-running frequency has been designed, providing a stable phase-shift variation at intermediate frequency of 450°.

References

[1] S. Ver Hoeye, A. Suárez, S. Sancho, "Analysis of Noise Effects on the Nonlinear Dynamics of Synchronized Oscillators", IEEE Microwave and Wireless Components Letters, vol. 11, no. 9, pp. 376-378, Sep. 2001.
[2] L.-F. Herrán, S. Ver Hoeye, F. Las Heras, "Nonlinear Optimization Tools for the Design of Microwave High-Conversion Gain Harmonic Self-Oscillating mixers", IEEE Microwave and Wireless Components Letters, vol. 16, no. 1, pp. 16-18, Jan. 2006.
[3] S. Ver Hoeye, L. Gutiérez, S. Sancho, A. Suárez, P. González, "Sub-harmonic and rational syncrhonization for phase-noise improvement," 31st Eur. Microwave Conf. London, U.K., September 24-28, 2001.
[4] E. de Cos, A. Suárez, S. Sancho, "Envelope Transient Analysis of Self-Oscillating Mixers," IEEE Transactions on Microwave Theory and Techniques, vol. 52, no. 4, Apr. 2004.

Figures

Reconfigurable and active integrated antennas have made rapid advances over the past decade, becoming popular in antenna engineering because of their frequency, radiation pattern and polarization diversity abilities. Hence, they are attractive for many military and commercial applications, such as radars, communication satellites, aircrafts, remote sensing, even electronic intelligence.

In most cases, electronic control is added to an antenna to achieve operation in a wider frequency band, turning on and off different parts of the antenna to modify the resonant frequency. However, in curved antenna array design, polarisation control is no less interesting. This is due to the fact that even if the fields radiated from individual elements in a curved antenna array are free of cross polarisation, the interference between fields radiated from different elements can in some directions increase the cross polarisation. The possibility to align polarisations of different radiating elements in an array could therefore significantly increase the power radiated in a given direction without increasing the overall power consumption, which would consequently increase the maximum scan angle of the array.

We propose a novel antenna with electronic polarisation switching that can be used as a basic building element in spherical (or curved in general) arrays. The antenna is an aperture coupled circular patch with three feeding slots separated by 120 degrees, enabling polarisation rotation in discrete steps of 60 degrees. The patch is fed from one centrally located coaxial connector, and the current is distributed to different excitation lines with RF switches. Such a geometry allows to maintain perfect symmetry and phase coherence between the polarisations.

The proposed architecture requires two sets of switches. One is located at the beginning of the feeding lines and is used to select the polarisation, while the other is used to additionally neutralise inactive slots (i.e. polarisations). Hence, lower mutual coupling and better polarisation purity are achieved.

It should be pointed out that the same architecture with one central coaxial feed can be used to achieve finer polarisation steps (e.g. 36 degrees, with 5 feeding slots), with only a moderate increase in the control circuit complexity. The antenna dimensions remain the same and therefore the proposed antenna is suitable for integration in large conformal arrays.

The rapid increase in the use of wireless communication systems is forcing more sophisticated base station technology to be developed. The agile beams of electronically scanned arrays provide significant system advantages; phased arrays technology is receiving considerable attention by the industry for ground base applications. This is accomplished using Butler matrix [1].

This study concentrates on the development of a switched beam antenna with cosecant squared elevation pattern in the European project BROADWAN framework. This antenna has simultaneously two functions.

The former is the beam scanning in azimuth direction. In order to accomplish this task, a 5 x 8 Butler matrix based on a novel broadside beam topologies [2] is presented and implemented at 42 GHz using microstrip technology. This Butler Matrix connected to an eight designed antenna arrays at 42 GHz will improve the performances of wireless networks by increasing the angular resolution of the base station antennas and avoiding a shadowed area in the broadside direction.

The latter is the shaping of a cosecant squared pattern in the elevation direction. This antenna is an array of eight printed patches with different weightings in order to concentrate the radiated energy below the horizon line and form the wished pattern. Thus, the illuminated sector is covered by a uniform power.

The antenna is printed on two distinct substrate layers with ground plane between them. The Butler matrix and the feed lines are isolated from the antenna arrays which are fed by slots in the ground plane to avoid the parasitic radiation.

The figures below show the layout of the whole system, the cosecant shaped radiation pattern in elevation plane provided by one sub array oriented in oy direction, and the steered beam in azimutal plane when the input 3 of the matrix is fed. The antenna has been fabricated to validate our simulations and the measure process is in progress.

References:

[1] J. BUTLER and R. LOWE, "Beam-Forming Matrix Simplifies Design of Electrically Scanned Antennas," Electronic Design, April 12, 1961

[2] M. KOUBEISSI, C. DECROZE, T. MONEDIERE, B. JECKO, "A New Method to Design a Butler Matrix with Broadside Beam: Application to a Multibeam Antenna"Microwave and Optical Technology Letters Vol. 48, Iss. 1, January 2006, Pages: 35-40

In the future radio communications, several radio interfaces operating at different frequency ranges must be integrated in a mobile device. For this reason, there is a growing demand for the frequency reconfigurable mobile terminal antennas. Different frequency ranges are needed in e.g. GSM, WLAN, 3G and especially in 4G telecommunication system, which is designed to utilize all the previously used radio and frequency resources.

Annular slot antennas are attractive choice for mobile applications because they are planar, lightweight, cost effective and easy to fabricate. Also its polarization and radiation pattern is able to fulfil the specification set to the mobile terminal antennas. In this paper, a frequency reconfigurable microstrip-fed annular slot antenna (Fig. 1a) is designed, fabricated and measured. The radius r and the width s of the slot are 21 mm and 2 mm, respectively. The prototype antennas are manufactured on Rogers RO4003 laminate with the relative permittivity, ε_r of 3.38.

Fig. 1. (a) Structure and (b) measured reflection coefficients of the annular slot antenna.

The frequency tuning is arranged with one RF PIN diode -switch (Infineo's BAR50-02V) located in the microstrip feed. Biasing the switch either by a forward or reverse DC-voltage, the length of the microstrip feed changes and the slot antenna resonates at different frequencies (Fig. 1b). When the PIN switch is reverse biased (V_bias = 1V), the length of the microstrip feed (AB in Fig. 1a) is 6 mm and the electromagnetic coupling from the microstrip to the resonating slot at 1.64 GHz results the lowest resonance. On the other hand, biasing the switch with a forward voltage (V_bias = +1V), the length of feed (AC in Fig. 1) is 23 mm and the resonance occurs at 2.31 GHz.

Using this configuration for a frequency tuning of an annular slot antenna by RF PIN diode and DC voltage, the frequency shift over 40 % is achieved. Additionally, the -10 dB bandwidths were reasonable being 8 % with the reverse biasing and 12 % with the forward biasing. The results also show that the proposed tuning method does not significantly decrease the efficiency of antenna when compared with the counterpart antenna without any tuning system.
In the final paper, a detailed antenna structure, and electric fields and surface currents around it will be presented. Furthermore, the paper will show the measured efficiency and radiation patterns of the antenna.

Antenna diversities have shown strong potential in wireless communications [1-3] and have been studied extensively in microwave range. Interest in reconfigurable antennas operating in millimeter wave range is currently emerging in various areas; typical planned applications include automotive radar, house automation... Most of time, the frequency and the polarization diversities are studied. Here, the radiation pattern diversity is overviewed. A directivity diversity antenna is helpful for cruise control radar application because the range detection can be adapted: a long range is possible with a high directivity, a short range with low directivity. So, a large environment in the front of the car can be analyzed.

This paper reports on a two microstrip reconfigurable antennas in millimeter wave ranges. An original method is outlined to guarantee matching for each configuration without changing circuit.

The objective for the Smart Tapering Active (STA) antenna is to have variable directivity. To that effect, the main idea is that the -3 dB beamwidth is proportional to the number of radiating elements. An appropriate architecture would be composed of n radiating elements and n switches. The number of ON-state switch(s) is varying. So, the feed network for the STA antenna is unconventional to guarantee matching for all configurations. A design procedure is reviewed in this paper.

To demonstrate the principles and capabilities of this STA antenna in millimeter wave, a 24-GHz and a 77-GHz four-element prototypes were fabricated. Great results have been achieved (Fig. 1-2) with prototypes based on 4-element array. The different configurations are named # n, where n is the number of ON-state switch(s). Obtained results are also validating the original method used to guarantee good matching for each configuration.

[1] W.H. Weedon, W.J. Payne, G.M. Rebeiz, J.S. Herd, and M. Champion, MEMS-Switched Reconfigurable Multi-Band Antenna : Design and Modeling, Proceedings of the 1999 Antenna Applications Symposium, Monticello 15-17 Sept. 1999.
[2] B. A. Cetiner, H. P. Chang, J. Y. Qian, M. Bachman, G.P. Li, and F. De Flaviis, Monolithic Integration of RF MEMS Switches With A Diversity Antenna on PCB Substrate, IEEE Trans. Microwave Theory and Tech., Vol. 51, No.1, January 2003.
[3] Darren S. Goshi, Yuanxun Wang, and Tatsuo Itoh, A Compact Digital Beamforming SMILE Array for Mobile Communications, IEEE Trans. Microwave Theory and Tech., Vol. 52, No. 12, December 2004.

A distributed antenna jointly processes the signals from widely separated antenna units. The antenna units may be connected to a processing centre by an overlay transmission network using various technologies, e.g. radio-over-fibre. This allows the distributed antenna to achieve gain not only at the link level but also at the system level. The distributed antenna can have an ability to dynamically reconfigure itself to preserve communication with users, particularly in mobile environments to combat the temporal nature of the environment.

It has been shown that for a distributed antenna base-station, mobile users have improved uplink SIR compared with that offered by a traditional, single antenna, base-station. Here we study the downlink user capacity of a distributed antenna cell using co-phasing transmission diversity. We construct the signal model for co-phasing transmission diversity and maximal ratio power allocation. Based on this signal model, we find a power control scheme by solving an eigenvalue problem. We then simulate the scheme to obtain the cumulative distribution of downlink SIR and user capacity.

Co-phasing transmission diversity can improve downlink SIR. It does this by aligning the impulse responses of the channels from all antenna units in time and co-phasing the peak components. Since there is typically modest phase variation of the components near the peak, co-phasing of the peak components will in many cases, result in most other components summing, approximately, constructively.

On the downlink, signals transmitted from multiple antenna units sum at a single receiving antenna, possibly destructively. This multiple-to-one topology requires a transmission diversity that can either isolate signals, and/or arrange for the constructive addition of signals, from all antenna units. Co-phase transmission diversity pre-adjusts the transmission delay and phase at every antenna unit such that the intended signal is combined constructively at the receiving antenna. One way to allocate power between antenna units is to follow the principles of maximal ratio combining. The principal feature of this scheme is that for each user the signals from all channels are summed in amplitude, whereas interference is summed power-wise.

Since the propagation distances from each antenna unit to the mobile users is different to that of other users, an equal share of total transmitted power at every unit cannot assure balanced SIR at every mobile user. Controlling the total transmitted power for each mobile user can, however, equalize the SIRs across all mobile users.

The figures show the SIR CDFs for 8 and 16 users. It is clear that with co-phasing diversity and the power allocation algorithm the distributed antenna achieves SIR gain. There is approximately 3dB gain from a single antenna unit to two antenna units and as more antennas are added the gain increases. Examining the CDF for each configuration, we find that the downlink SIR for the distributed antenna is not constant. The power control algorithm does equalize SIR for all users. The SIR resulting from power control is not the same for different mobile user spatial distributions. The gradient of the CDF is related to the number of antenna units, the more antenna units, the smaller the slope. This is because when more antenna units are deployed, the number of ways of clustering the mobile users around these units increase.

In this paper, a multibeam and orthogonal polarization antenna is considered in order to improve performances of ad-hoc networks as well as MIMO systems. Ad-hoc networks are wireless point to point systems. No base station is needed, each communicant terminal can have such a function. Of course, in this multi-users environment, signals are mutually disturbed. Furthermore, since the communicating nodes are mobile, the network topology is continuously changing. Taking into account these problems, some specified signal processing algorithms have been developed. The possibility of deferring part of signal processing constraints on the antenna in order to maximise the transmission quality is the main idea develop in this paper.

In such systems, reconfigurable antennas offer a new dimension which can be used to enhance communication performance. In [1], the author studies the efficiency of a network using smart antennas. The beamforming is used to control the main direction of propagation and to place nulls toward the interfering signals direction. So the signal to interference plus noise ratio can be improved. Nevertheless, this technique is limited in a few range of aperture angle in the case of a classic antenna array use and the beamforming control complexity is relatively high.

In indoor communications, some multipath exist involving fading effects from some direction. That is why our antenna has the capability of switching its main lobe on 4 pi steradians range.

An other way to improve the transmission capacity is to use this antenna as a single element in MIMO system. Indeed, it produces three orthogonal polarizations, each on different directions. So it will be less sensitive to the null direction effect.

The antenna design is based on a metallic cube full of air. On each faces, a rectangular slot is etched. These slots are fed by a monopole penetrating the cube through one of its corner. This structure presents three different resonances due to the monopole, to the cavity and to the slots. Because of its geometry, three orthogonal polarizations are performed by the cube antenna. By short-cutting slots, it is possible to null the radiation in desired directions.

In the full version of the paper, the authors will present a new simple way of improving communication performance using a reconfigurable multibeam and orthogonal polarizations antenna. After considering the antenna mechanism, simulation and realization results will be presented and compared.

[1] S. Bellofiore and al., "Smart antenna system analysis, integration and performance for mobile ad-hoc networks (MANETs)", IEEE Trans. Antennas and Propagation, vol. 50, n°2, May 2002.

Wireless telecommunication devices like mobile phones today utilize a growing number of telecommunication standards operating different frequency bands. To decrease the amount of antennas in a same device and to enable further miniaturization, there is a demand for compact efficient antennas having tunable operation frequency with maintained radiation characteristics.

In this paper, two frequency reconfigurable planar inverted F-antennas (PIFAs) applicable for portable wireless devices are studied. The antennas are designed to operate around 1.8 GHz, fabricated on Rogers R4003 laminate and validated with measurements. Surface mount semiconductor RF-switch components are incorporated to the antennas to implement the electrical frequency tuning. Two different approaches are used to enable frequency tuning. In the first PIFA structure (Fig. 1a), the frequency shift is achieved by modifying the resonating current path of the antenna with RF PIN diodes. In the second one (Fig. 2a), the operation frequency is tuned with a SPDT RF CMOS switch by loading the antenna with either a short circuited or an open ended tuning stub.

The frequency shift, referred to the center frequency, for the first and second PIFA were 20 % and 6.5 %, respectively (Fig. 1b and 2b). Both antennas with the tuning circuits maintained broadband impedance characteristics with -10 dB bandwidth more than 8 % and 13 %, respectively. Furthermore, the antennas proved to be effective radiators with almost omni directional radiation patterns. The measured radiation patterns and total antenna efficiencies are presented and discussed in the final paper.

Keywords: Reconfigurable antenna, PIFA, RF switch

Fig.1. (a) Photograph and (b) measured return loss of the reconfigurable PIFA with RF diodes.

Fig.2. (a) Photograph and (b) measured return loss of the reconfigurable PIFA with RF CMOS switch.

The target detection with monopulse radar antennas requires the generation of sum and difference patterns at the same time. Necessarily, some design constraints have to be taken into account to guarantee low side lobe levels, high directivity and narrow beam-width. In such a framework, analytical methods have been proposed for designing independently optimal excitation coefficients for the sum [1] and the difference [2] pattern. Unfortunately, such approaches require the implementation of two independent feed networks that is usually expensive. Therefore, there is the need of finding some compromises in the optimal design of sum and difference patterns in order to reduce the costs of the antenna system. A common strategy employed in the antenna synthesis is that of firstly computing the optimal excitation coefficients for the sum pattern and successively to determine the weights of suitable sub-arrays in order to generate an approximate difference pattern. Consequently, two sets of unknowns are taken into account: the aggregations of the N elements of the array in Q sub-arrays and the weights of each sub-array. Such a problem has been addressed in the state-of-the-art literature by means of analytical methods [3] or optimization procedures [4][5].

The approaches proposed in this work take into account that the partition of the elements of the array can be carried out considering those having similar properties. Towards this aim, the residual error sorting method (RES) and the gain sorting method (GS) have been developed in order to group elements with similar optimal residual error and with similar optimal gain [3], respectively. The solutions space is sampled by covering a non-complete binary tree for finding the minimal cost path from the root to the leaves so that the array configuration minimizes a suitable cost function. Towards this end, the solution is looked for by considering some special elements inside the aggregations called border elements. These elements can be moved between closed sub-arrays without violating the sorting property. Such techniques allow the reduction of the investigation space and a significant computational saving as well as an enhanced accuracy in satisfying the user-defined constraints for practical applications.

References

[1] T. T. Taylor, "Design of a Line-Source Antennas for Narrow Beam-Width and Low Side Lobes," Trans. IRE, vol. AP-7, pp. 16-28, Jan. 1955.
[2] D. A. McNamara, "Discrete n-distributions for difference pattern", Electron. Lett., vol. 22, pp. 303-304, Jun. 1986.
[3] D. A. McNamara, "Synthesis of sub-arrayed monopulse linear arrays through matching of independently optimum sum and difference excitations," IEEE Proc. H, vol. 135, pp. 371-374, Oct. 1988.
[4] F. Ares, S. R. Rengarajan, J. A. Rodriguez, and E. Moreno, "Optimal Compromise Among Sum and Difference Patterns Through Sub-Arraying," IEEE Antennas Propagat. Society International Symposium, pp. 1142-1145, July 1996.
[5] P.Lopez, J. A. Rodriguez, F. Ares, and E. Moreno, "Subarray weighting for Difference Patterns of Monopulse Antennas: Joint Optimization of Subarray Configurations and Weights," IEEE Trans. Antennas Propagat., vol. 49, pp 1606-1608, Nov. 2001.

An example corresponding to future European coverage scenario for satellite with frequency reuse is presented in Fig. 1a. The polygons surrounded by the same type of line (A, B and F; then C and D; then E and I; finally, G and H) must be illuminated by the same frequency, while maintaining between their beams an isolation of at least 27dBi (for example, the beam centered on E must have on I a maximum gain attenuated of 27 dBi compared to the minimal gain on E). Balling et al.

[1] use a configuration made up of parabolic reflectors illuminated by a huge number of feed elements (90 on the whole). This solution is complicated to implement, because of the Beam Forming Network (BFN) used to control the radiation diagram. In this paper, we chose to work with a configuration made up of shaped reflectors to control the radiation diagram, illuminated by a few number of feed elements, which radically reduces the BFN complexity: for example, E and I are illuminated by only two feeders and one shaped reflector. On the other hand, 120 degrees of freedom are used for the synthesis of this shaped reflector (defined by a modify Jacobi development [2]).

The design of this new solution is a difficult optimization problem, with a huge number of design parameters, some constraints and several competing objectives: typically, we have to maximize the minimal directive gains for the principal polarization on each zone, and the isolation between them. We propose to use the Multi-objective Genetic Algorithm

[3], which does not require any gradient information. Fig. 1.b. and 1.c. show the results obtained by our methodology.

(a) Possible European coverage scenario with the frequency reuse

(b) E area coverage

(c) I area coverage
Fig 1. Contour Beam Shaped Reflector Antenna with Frequency Reuse

References

[1] P. Balling et al., "Design and analysis of large linearly polarised array-fed offset reflector antennas with frequency reuse," In: 26th ESA Antenna Technology Workshop on Satellite, The Netherlands, Nov. 2003.

[2] D.W. Duan and Y. Rahmat-Samii, "A Generalized Diffraction Synthesis Technique for High Performance Reflector Antennas," IEEE - Antennas and Propagation Magazine, v. 43, n. 1, pp. 27-40, Jan. 1995.

[3] C. A. Coello Coello, List of references on evolutionary multi-objective optimization, [online]. http://delta.cs.cinvestav.mx/~ccoello/EMOO/, 2006.

I. Introduction

New wireless applications involving small wireless devices raise the problem of antenna non isotropy. Ad hoc sensor networks monitoring, human motion capture systems as well as localization of mobile objects are typical upcoming applications demanding for highly reliable transmission performance whatever are orientations of the devices relative to each other. Because of sensors small dimensions, multi port antennas with only few elements can be integrated and have to be miniaturized. Due to low power and low cost requirement, antenna processing has to be reduced to the minimum. This give rise to a need for small antennas optimized to achieve full isotropic dual polarization coverage with low complexity antenna processing. In many ad hoc wireless network applications, distances between neighbour nodes are short or unobstructed. The line-of-sight (LOS) path is predominant and fading occurs because of antenna radiating hole or polarization mismatch. Our approach is to compute the coverage of multi elements antennas in free space for all of the LOS path directions of arrival (DoA) and all of its polarization tilt. Based on this method, we propose a comparative study of the coverage performances of five canonical multi port antennas.

II. Coverage principle

The characterization method [1] will be rigorously presented in the final paper. The multi port antenna is supposed in reception mode in free space. It is illuminated by an incident plane wave (IPW). Signals induced at each port are computed and the strongest one is chosen (selection combining). The coverage is then computed as the proportion such that the received power is above a given threshold when directions and polarization tilt vary. Results are given as plots relating the coverage whatever is the direction or the orientation of the antenna relative to the transmitter as a function of the minimum received power requirement. In plots shown below the received power is normalized to the mean radiating power and thus represents antennas directivity.

III. Comparison of the multi elements antennas

Five canonical radiating structures have been compared: A single dipole, Two cross-dipoles, Two U-shaped bended cross-dipoles, A combination of a dipole and a loop, Three orthogonal dipoles. Two plots are presented in fig.1: (a) for a linearly polarized IPW and (b) for a circularly polarized IPW. The single dipole and the three orthogonal dipoles give lower and upper bound coverage. Comparison of the two ports structures will be commented in the final paper.

IV. Conclusion

In the final paper, results will be commented and completed with more realistic antennas. A comparison of differents combining methods will be detailed. The present study intends to give a novel approach for optimization of practical multi antennas structures in order to achieve full isotropic coverage and an associated characterization tool.

V. References

[1] M.Huchard, C. Delaveaud, S. Tedjini, "Characterization of the Coverage Uniformity of an Antenna based on its Far-Field", IEEE AP-S 2005, Vol. 1B, pp 500 -503, 2005

The paper describes a conception of a linear microstrip antenna array, which is capable to transmit and receive several variants of linearly polarized EM waves at GSM1800 frequency band (i.e. slant, vertical and horizontal polarization). The basic idea which stands behind selection of the antenna structure is a possibility for simultaneous and independent transmitting and receiving two mutually orthogonal slant polarizations (i.e. +/-45°) with high level of isolation between them (higher than 30dB).

The first part of the antenna project was concerned with the design of a single radiating element. In order to meet requirements for its broadband performance (about 10%) we decided to use the conventional aperture coupled microstrip patch antenna, additionally equipped with a back ground plane (Fig. 1).

In order to achieve the assumed polarization characteristics, two orthogonal slant polarizations were obtained by means of two radiating elements, which were rotated suitably by the angle of +45°and -45°in relation to vertical polarization. A couple of such elements where used to built the antenna array, so the whole antenna structure is composed of two linear arrays, and each of them contains four radiating elements. The antenna array structure is shown in Fig. 2. The overall size of the antenna is 265x458mm (DxL). All radiating elements are fed by means of T-junction based corporate feed network, which provides uniform power distribution.

Preliminary evaluation (computer simulation) shows that the proposed structure works in the required frequency band with the return loss less than -16dB. The isolation between the two input ports is better than 30dB and for some frequencies even as high as 38dB. The full paper shows results of measurement for the antenna model, which was built in the experimental stage of research. The results focus, in particular, on the antenna radiation patterns measured at different polarizations.

The natural choice of polarisation for a Wireless Sensor Network (WSN) is not obvious, and the IEEE 802.11x standards at present contain no recommendation. It was suggested that horizontal polarisation [1] may reflect more strongly from ceilings, while there is also the possibility of using switched - polarisation diversity to improve reliability. Likewise it may be possible to improve frequency re-use by using orthogonal polarisations to increase isolation or using dual-polarized form of ring array by exciting square patches in two orthogonal modes [2]. This given uncertainties about the propagation conditions, it would be useful to have a free choice of polarisation, and a polarisation diversity scheme might also be considered. An interesting problem for such wireless networks is the use of conical beam radiators that consider the peak radiation could be made to occur at higher values of theta, i.e. lower elevations. This also explains the type of mounting of these antennas on ceilings and ground surface.

This paper presents two circularly-polarised conical beam antennas for short-range low power Wireless Sensor Network application, such as Bluetooth, WiFi and ZigBee. This work represents an extension of the author research in [2] that used of the horizontal or vertical polarised conical beams antenna arrays. The antenna was also realised as a group of microstrip patch radiators in a ring formation as shown in Fig. 1 and 2. Three and four elements antenna arrays for circularly-polarized were designed and discussed for small elevation angles. Two well known different feeding types of these antennas were also considered.

References:

[1] Y.J. Guo, A. Paez, R.A. Sadeghzadeh and S.K. Barton, "A circular patch antenna for radio LANs", IEEE Trans. Antennas and Propagation, Vol. 2, pp. 177 - 178, 1997.
[2] N.J.McEwan, R.A.A.Alhameed, E.M.Ibrahim, P.S.Excell and J.G.Gadiner, "A new design of horizontally polarised and dual-polarised uniplanar conical beam antennas for hiperlan", IEEE Trans. Ant. And Propogat., vol.51, Feb.2003, pp.229-237