When a high gain antenna for a high frequency system is being designed, low losses feeding networks must be taken into account. To show how parallel plate feeding networks have been used a planar array antenna working at the 37 GHz band will be presented.

The system requires right hand side polarization (RHCP) with axial ratio under 3dB and the radiating element used (modified circular patch) provides it. To couple the power from the parallel plate waveguide to the radiating element, properly designed microstrip coupling lines will be used. The design is a monopulse bidimensional patch array integrated in a narrow band application (BW<1%) for signal detection. The specifications in terms of radiation pattern are: monopulse in the horizontal plane with low side lobes (20 dB below the main lobe) in the sum pattern (Figure 1); the difference radiation pattern must be over the sum radiation pattern except in an angular region of 6° located in the centre of both patterns; cosecant shape radiation pattern in the vertical plane (Figure 2). The obtained gain must be greater than 32.5 dB. Monopulse working implies a deep study of the fields generated in the parallel plate.

The structure present three different modules, namely: the radiating circuit (patches and microstrip coupling lines), the parallel plate waveguide and the network that couples power to the parallel plate.

Figure 3 shows a transverse view of the structure. The radiating circuit consists of two equal substrate layers (dielectric constant 2.17), in the lower one the microstrip coupling lines are printed and the patches are printed in the upper one. Microstrip coupling lines are in charge of coupling power from the parallel plate waveguide to the radiating elements through a metallic via hole that connects them. The dimensions of the coupling lines control the amplitude of the power taken to the radiating elements following some feeding function to obtain the required radiation pattern. The phase control of the radiating element is carried out by the rotation of the element around its feeding point. The metallic layer in the middle of both substrates has two functions: ground plane for the patches and the microstrip lines, and the cover of the feeding structure that totally closes the parallel plate waveguide. To work with the fundamental mode (TEM) a distance lower to lg/2 between the parallel plate is required.

To feed the parallel plate waveguide a probe feed network connected to rectangular waveguides has been employed. To generate a monopulse in the horizontal plane two probes are needed, so if the sum radiation pattern is required both inputs have to be excited with the same phase. If the difference radiation pattern is needed, a phase difference of 180° between the inputs must be forced. If a monopulse in both vertical and horizontal planes is required, four inputs will have to be used to feed the waveguide with the appropriate phase distribution. In any of the mentioned cases it is necessary to minimize the active reflection in every single input and for each working mode (sum or difference), that is the reason why a matching pin is used.

Topic: Onboard antennas
Branko M. Kolundzija, Dept. of EE, University of Belgrade, PO Box 35-54, 11120 Belgrade, Serbia and Montenegro, e-mail: kol@etf.bg.ac.yu, phone: +381113218329, fax: +381113015655
Contact person: Branko Kolundzija

Simulation of antennas placed on large platforms is computationally very demanding. Electrical size of the structures imposes the use of special techniques for simulation on PCs. Method of moments analysis based on surface integral equations using lowest order basis functions and triangular mesh leads to a large number of unknowns, e.g. 150 per wavelength squared for metallic structures. With the application of higher order basis functions, modeling is much more efficient, with 30 unknowns per wavelength squared. However, for electrically very large structures, simulation on a PC is still tedious.

By reducing the order of current expansion all over the structure, a large reduction of the number of unknowns is achieved, but with significant loss of accuracy. What is proposed here is to adaptively decrease the order of expansion in the regions of the platform which are distant from the antenna or shadowed. This allows significant reduction of the number of unknowns while preserving good accuracy of calculated radiation pattern or coupling between multiple antennas.

In the formulation used by WIPL-D 3D EM solver, the platform is meshed into quadrilateral patches. The order of expansion defined over a patch is dependent on the electrical size of the patch. If the longest patch side exceeds 1/6 of the wavelength, second order is used. For each additional 1/3 of the wavelength, the order is increased by one. Let’s define a reduction coefficient C which will multiply the order of expansion obtained in the aforesaid way. For a single antenna, C is given with 1-k(r/rmax), where |k|<1, r is the distance from the antenna to the observed patch, and rmax is the distance from the antenna to the farthest patch. In case of two antennas, C is given as 1-k(r1+r2-d)/((r1+r2)max-d), where r1 and r2 are distances from the first and second antenna to the observed patch, (r1+r2)max is the maximum sum of r1 and r2, and d is the distance between antennas. Both cases are shown in Figs. 1 and 2.

In the previous equations, k is varied in the range 0.2-0.8 which influences the steepness of reduction of expansion order. Model shown here is an ellipsoid with axes equal to 10 wavelengths (x-axis and z-axis), and 20 wavelengths (y-axis), with a 1/4 wavelength monopole placed on the surface along the x-axis. The half of the ellipsoid opposite to the antenna is considered a shadow, with C equal to 0.1. Results for Boeing payload fairing model will be shown in the paper.

When the reduction is performed linearly on the entire platform, radiation pattern results vary a lot as the number of unknowns decreases (Fig. 3). When the adaptive reduction is performed, variation of the radiation pattern results is much smaller, while the number of unknowns is reduced dramatically (Fig. 4). In this case, adaptive reduction provides acceptable results at about 4 times less unknowns then the original expansions.

Fig. 1 Fig. 2
Fig. 3 Fig. 4

SAR antennas on satellites are usually arrays. Missions like Envisat and TerraSAR-X uses active phased arrays with large flexibility and wide swath coverage. The active array systems are however both expensive and heavy. At the other end of the scale there is the single beam reflector antenna. The obvious advantage with these systems is the low weight, large bandwidth and low cost. Even relatively large reflectors can be launched with low cost launchers and the required deployment can be made with the technology used for telecom antennas. A SAR system based on a reflector antenna can give a high resolution at low cost on the expense of flexibility and swath coverage. This paper describes a feed system for a reflector antenna that will give multiple dual polarized beams, combining some of the flexibility and wide swath coverage of the phased array with the low cost and weight of the reflector system.

The principles and performance of such system has been investigated in an ESA study running in 2004 by EADS Astrium GmbH and Saab Ericsson Space AB. The study recommended an offset elliptical reflector antenna fed by a linear array, "the multi pencil beam concept". In 2005 the same companies were awarded a continuation of the study by ESA. The continued study involves design, manufacturing and RF-testing of a feed array breadboard. The paper focuses on the requirement and the design of the feed array. The design must be multipaction free at high peak power. The feed array shall also give the same elliptical illumination for both vertical and horizontal polarization. The proposed feed array consists of a linear array of "split horns". Each horn consists of two apertures to avoid grating lobes due to the large element separation. The horn aperture are flared and corrugated in the dimension perpendicular to the linear array and has a constant width along the linear array. The corrugations give the same E and H polarized pattern in the plane perpendicular to the array and the element pattern along the array is dominated by the array factor. The feed array is currently under manufacturing and the RF measurement of the array will be presented.

Tapered-slot antennas became popular in the last decade in different applications such as communications, ground penetrating radar (GPR) etc. They are wideband, low-dispersive (thus, do not distort short pulses considerably), light-weight and low-cost. Several types of baluns that work as a wideband transition between asymmetric microstrip or coplanar line and symmetric slotline had been developed. Besides, in GPR applications signals of tapered-slot antennas are not much affected by the vicinity of the ground as is the case e.g. in waveguide horn and bow-tie antennas. Nevertheless, flare ends of the tapered-slot antenna behave as discontinuities deteriorating the antenna performance. So, multiple reflections between balun and flare ends cause ripple in gain as a function of frequency. In GPR, interactions between flare ends and ground surface produce clutter in shallow region that can obscure shallowly buried targets. That is why several designs had been proposed to match flare ends especially at low frequencies, and those are as follows.

Multihead Configuration for Ground Penetrating Radar and Depth GPR systems can be configured in a variety of ways in order to optimise their performance. In this paper multihead GPR systems are examined to establish the best configuration in terms of determining the depth of a target. The analysis examines the errors that may be introduced when a layered sub-surface structure, as is commonly found in urban streets, is treated as a single infill.

AThe challenge is to accurately resolve the positions of buried objects, not only in their plan position but also their depth. Indeed the Mapping the Underworld project aims to find target positions to an accuracy of 3cm or better. This suggests increasing the bandwidth of the radar system to improve the temporal resolution. However, increasing the bandwidth naturally leads to the use of higher frequencies where the attenuation characteristics of typical soils are much more. With this in mind a multihead system is considered with a view to establishing how to best configure the system for best accuracy in target depth.

When using a GPR system the permittivity of the ground in which the target is buried is not known. The general configuration of a multihead GPR system is shown in the diagram. Analysing the propagation time delays between the heads and the target positions yields several measurements with common factors of target depth and the permittivity of the infill. These can be manipulated to eliminate the unknown permittivity of the infill and determine the depth of the target.

Measurements are always prone to some degree of uncertainty, and by performing a sensitivity analysis on the equations for target depth, the sensitivity of various head configurations to uncertainties in measured time delays can be examined. The results of this are shown in the diagram. It is apparent that care needs to be taken in choosing the best configuration of the transmitter and receiver positions.

Many configurations show a high uncertainty for shallow targets where g ~ 0. All configurations show minimum uncertainty when the inter-antenna spacing is about the same, g ~1, as the depth of the target. For targets at a depth of 2m the 5 head system would be 10m wide, which is obviously too wide for mapping targets below a road or pavement. For use across pavements a spacing of S=0.25m and a total width of 1.25m is far more realistic. In that case a target at a depth of 2 m corresponds to g=8. Clearly to achieve minimum uncertainty up to such depths the best configuration is to use paths B to A and B to C, where it is better than the other configurations by a factor of 6 or 7.

An FMCW radar system has been built using these principles and measurements taken over an in-house test sand pit. These have shown that the depths of 3 relatively closely spaced targets were determined to accuracies of a few centimetres for targets at depths around 15cm to 30cm. The reported permittivity of the sand infill is ε_r~3 which agrees well with published data for fairly wet sand.

In order to resolve closely spaced targets in a planar surface normal to the beam, the antenna used in the radar assessment of concrete structures should have a narrow beam width which means a higher directivity. This work presents a procedure to optimize the field pattern of a bow-tie antenna using multi-objective genetic algorithms [1] in conjunction with a MoM direct solver.

This procedure searches to attend two objectives: to minimize the metal area of the antenna (in order to reduce the size) and to maximize the gain in the plane normal to the antenna. The moment method (MoM) with Rao-Wilton-Glisson (RWG) basis functions [2] is used to calculate the electromagnetic characteristics of the antenna.

In order to find the antenna configuration with a higher directivity and a smaller metal area we implemented a multi-objective genetic algorithm to accomplish two conflicting objectives. The parameters to be adjusted by the MGA are the variables necessary to obtain the far-field characteristics of the antenna in the air: the length L [0.1ë to 1ë] (with frequency equal to 1GHz), the flare angle α =[30° to 120°], and the void spaces in the antenna structure.

Fig 1 shows the modifications applied to a bow-tie antenna in order to improve the radiation pattern. The optimized antenna proposed by the algorithm has α=79° and L=26cm with 11% of the elements erased. The gain obtained in the plan normal to the antenna was 6.37 dB against 3.40 dB of a common structure with improvement of the half-power beam width from 57.6° to 43.2°.

The significance of the antenna pattern optimization is in the resolution that can be achieved in improving the antenna design. The results show that a better field pattern can be obtained with the optimized antenna which leads to a better signal penetration and more realistic GPR images of lossy concrete structures.

[1] K. Deb, Multi-Objective Optimization. John Wiley & Sons, 2002.

[2] S. Makarov. "MoM antenna simulations, with Matlab: RWG basis functions," IEEE Antennas and Propagation Magazine, Vol. 43, No 5, pp. 100-107, October 2001.

The developpement of multisensor for ground penetrating radar (GPR) is slowed down by some constraints and difficulties that concern the complexity of electronic control system, setting antennas in presence of various ground, real time with realistic rendering of the electromagnetic signal from subsurface, robustness... Electromagnetic modelisation tools of multisensors in presence of realistic environnement can contribute to reduce the cost of experimental device by carrying out accurate information on the antennas behaviour, parasiting coupling and prediction of B-scan and C-scan.

A knowledge of 3D representation is crucial to achieve the accurate subsurface identification. In order to obtain these realistic C-scan, we model by FDTD the complete scene with buried objects of dielectrics or dispersive dielectric materials. The total scene is cuting out in computational volume according to x direction (with a dx step) and in the y direction (with a dy step). x and y are the azimuth direction and z the normal direction to the ground surface.This truncation brings a strong reduction of computatinal time. We thus obtain Nx volume according to x and Ny volumes according to y. Each subvolume is compared to a recording space Ground Penetrating Radar (GPR). We use Wu-King antenna model in our application but various antennas or network could be considered. Their Ultra-Wideband (UWB), their compactness, and the fact that they do not have cross polarisations make them well suited for GPR. The medium describing the ground can be based on the fractal model [1]. These model will be presented in the full paper.

We get a table containing the values of current according to time for the various positions (x, y) of the radar. By collecting all the results, we obtain the values of I (x, y, t) that give us the 3D representation of the total scene. In the example given, the scene measures 3m of length, 2m of width and 1m of depth. It is cut out all the 5 cm in x direction and 5 cm in y direction. Each subvolume measures 1,5m*1,5m*1m that give us 341 simulations. The truncated scene, with electrical wall boundary condition could reduce the duration of simulation under 8 minutes and approximately 45 hours for one C-scan. It is thus necessary to use several PC or a supercomputer to accomplish calculation quickly. We wish to use this methodology to simulate the performances of multisensor radars. The simulation data will be used to test migration algorithm and in particular reverse time migration algorithm [2] with the advantage of full controling the environnement and the multisensor configuration.

[1] Modélisation de milieux granuleux à l'aide de la mèthode dite « Diamond-Square4D »- Application à la simulation de radars GPR par FDTD, S. Besse, C. Guiffaut, A. Reineix [2] A matched-filter-based reverse-time migration algorithm for ground-penetrating radar data, C. Leuschen, R.G. Plumb

This paper deals with a first investigation on critical issues in the design of a Ku-band radiometer for fire detection. This activity is addressed in the framework of a national research program, sponsored by the Italian Ministry of University. Passive radiometers measure the intensity of radiation, and have been widely employed on satellite/aircraft installations for remote sensing of the earth. Here, we are interested in a ground based application for detection and monitoring of wildfires [T. Kaiser, T. Kempka, 2001][L.S. Sadovnik, V.A. Manasson, M. Mino, V. Kiseliov, 1999].

The antenna design plays a key role. Its internal temperature and directivity pattern drastically affect the sensitivity of the receiver front-end. In particular, maximum directivity, narrow beamwidth and low side lobe level (SLL) are desirable features, in order to reduce unwanted signals, relaxing thus the Noise Figure (NF) of the Low Noise Amplifier (LNA). Recently, microstrip array antennas have been investigated as good candidates to accomplish the above requirements. Thus, in order to get low cost and weight, and simplicity of manufacturing, as requested in our research program, the performances of a patch array antenna, working at 13 GHz with 200MHz bandwidth, are here investigated. As a first step, the antenna spec's have been roughly estimated accounting for several possible wildfire scenarios, where the main parameters are the fire distance, extension and temperature. For a detection range of hundred meters, a preliminary investigation suggests that a directivity of around 25dB forces a NF lower than 2dB, and an available power gain higher than 20dB for the LNA. These performances can be obtained with the most recent CMOS technologies, allowing an implementation of the whole radiometer on a single silicon chip. The activity has started by finding a first simple solution to reach the appropriate directivity for quite long distance detections (main task). A simple uniform, single slab, patch array antenna has been designed, prototyped and measured to reach at least a directivity of 25dB. Starting from this very first prototype, the aperture coupled technique has been then employed, in order reduce cross-polarization. The layout of each radiating element has been chosen as a multilayered structure, where standard dielectrics have been adopted [F. Rostan, W. Wiesbeck, J. J. van Zyl, 1996]. Further, because the antenna will be pointing towards the horizon, and placed relatively close to the ground, high SLL will be required, in order to diminish the radiation picked up through minor lobes from unwanted "hot" signals [D.J. McLaughlin, R.E. McIntosh, A. Pazmany, L. Hevizi, E. Boltniew, 1991][K.S. Kona, Y. Rahmat-Samii, 2005]. Then, the goal of the project will be to increase the beam efficiency, i.e. the power received through the main lobe with respect to the total received power, with choice of appropriate tapered feedings, still providing good directivity and simplicity in realization. A class of results, still under investigation, will be exposed at the conference.

Extensive work has been performed at JPL on the use of a Deformable Flat Plate (DFP) to correct for the gravity-induced distortions on a large reflector antenna. The DFP is placed in the beam path and deformed in order to compensate for the gravity-induced distortions as the antenna moves in elevation. Actuators controlling the plate surface are driven via a look-up table. Values in the look-up table are derived using the measured antenna distortions, ray tracing, and a structural finite element model of the DFP.

There is a proposal to apply the DFP concept to compensate for reflector distortions in Spacecraft Antennas. However, instead of a mechanically deformed plate, the wave front compensation is accomplished by using MEMS switches integrated with patch reflect array elements to change the phase (path length) of the reflected signal.

Nonetheless, for spacecraft antennas where the distortions can be due to imperfections in the manufacturing process, on orbit errors due to thermal effects or long term changes in material properties, the actual surface distortion is unknown and needs to be determined in order to provide the necessary compensation.

The purpose of this paper is to discuss algorithms for estimating the reflector distortions.

One algorithm explores the possibility of using the amplitude and phase of the signals received from an array of feeds in the focal plane. A simple statement of the problem is; given the relative amplitude and phase of the signals received by an array feed in a distorted reflector, compute the reflector distortions required to produce the given signals. The problem is solved in two parts. First, the focal plane field is estimated by best fitting a postulated focal plane field with the given signals. Then, the focal plane fields are used to compute the distortions. A second technique explores the possibility of an optical surface measuring system.

The SKA, or Square Kilometre Array telescope, is an international 'next-generation' radio telescope [1]. In Australia, CSIRO is working on a technology demonstrator for the SKA. The first stage is known as the "new technology demonstrator" or NTD. This is a two-antenna interferometer, located at CSIRO's Radiophysics Laboratory at Marsfield and aimed for completion in mid-2007. A key goal for the SKA is a large field of view (FoV), and the NTD aims to demonstrate large bandwidth and large FoV using small (14m) dishes equipped with wideband active phased-array feeds at the prime focus. The second stage, known as the "extended new technology demonstrator" or xNTD will use technology developed under stage one to build a working radiotelescope located in a radio-quiet zone at Mileura, Western Australia. The working specifications for the xNTD are:

The growing interest on high-speed Internet access for residential and small business costumers has created a demand for last mile broadband access. In this scenario, when compared with wired technologies, wireless technologies show lower costs, rapid deployment, and lower maintenance. Among the wireless services, LMDS offers a method of access to broadband interactive services, where a base station communicates with many fixed subscribes. Wireless links at millimeter waves make attractive the use of compact reflector antennas, capable of providing wider absolute bandwidths necessary to transmit wideband signals. Single and dual reflector antennas have been investigated for omnidirectional coverage. The reflector surfaces are bodies of revolution obtained by spinning confocal conic sections or shaped generating curves about a symmetry axis [1]-[5] (see Figure 1). As demonstrated in [3], the dual-reflector antenna leads to more compact designs than the single reflector concept, as the dual configuration requires considerably smaller reflector diameter to achieve the necessary aperture width for adequate control of the vertical radiation pattern. Besides the compactness, the use of axis-displaced dual-reflector configurations offers the additional advantage of controlling the feed return loss by minimizing the subreflector radiation towards the feed horn, as discussed in [5]. For efficient coverage of some LMDS scenarios it may be desirable a radiation pattern with main beam direction with its peak placed below the horizon to avoid interference [3],[4]. In this paper, we explore axis-displaced dual-reflector configurations to provide omnidirectional coverage with an arbitrary main-beam direction in the elevation plane, thus extending the investigations conducted in [5]. By applying GO principles, close-form equations for the classical dual-reflector antenna design are derived. Four different antenna configurations are identified and distinguished by two different options for the ray mapping and by the location of the subreflector caustics [5]. Figure 2 shows a geometry option with a subreflector virtual caustic. The design procedure is exemplified by applying the formulation in the synthesis of omnidirectional dual-reflector arrangements. The case studies explore the potentialities of these configurations as base-station antennas in point-multipoint radio links with a tilted main lobe. In order to estimate the limits for applicability of the proposed design procedure, the antennas are analyzed by the Method of Moments (MoM) technique. Figure 3 shows the MoM radiation patterns obtained by antennas with main beam directions at +/- 8 degrees with respect to the horizon.
ReferEncias:
[1] A. P. Norris and W. D. Waddoup, Proc.4th ICAP, 1985, pp. 141-145. [2] M. Orefice and P. Pirinoli, Electron. Lett., vol. 29, no. 25, pp. 2158-2159, Dec. 9, 1993. [3] A.. Pino, A. Acuña, and J. Lopez, Microw. Opt. Tech. Lett., vol. 27, no. 5, pp. 371-374,Dec. 5, 2000. [4] J. Bergmann, F. Hasselmann, and M. Branco, Microwave Opt. Tech. Lett., vol. 24,no. 6, pp. 426-429, March 20, 2000. [5] F. Moreira and J. Bergmann, IEEE AP-Trans.,vol. 53, no. 9, pp. 2799-2808, Sep. 2005.

Given the market`s economic and strategic constraints, communication satellite missions require high performance solutions with low cost and low risk.

In association with antenna designers, the search of both technical and economic performances triggered the development of the Ultra-Light Reflector concept by EADS Space, with the objective to offer a quasi-recurrent product, though easily adaptable to specific missions. The development of the first generation of such reflectors started in 1998. Today, the second generation of Ultra-Light Reflectors is a commercial success, as shown by the large family of reflectors delivered in the range of 2 to 4 meters aperture diameter.

First generation Ultra-Light Reflector

Second generation Ultra-Light Reflector featuring high-temperature shell technology and improved performances

Commercial Ultra-Light Reflector family

In the future, the performances are expected to be further improved by building on this heritage. This can be achieved by limiting non-standard reflector components to the bare essentials that are the shell shaping and the deployment interface. With such standardisation level, a fully modular justification philosophy can be proposed, while complying with the customer’s confidence and satisfaction.

A hologram can be used as a collimating element in a compact antenna test range (CATR) at millimetre and sub-millimetre wavelengths [1]-[3]. The performance of a hologram can be improved by modifying the illumination of the hologram with a dual reflector feed system (DRFS) [4]. A 650 GHz dual reflector feed system (DRFS) was designed as part of an ESA project aiming at the measurement of a 1.5 m antenna at 650 GHz during autumn 2006. Design of the 650 GHz DRFS is presented and simulation results are presented and compared to the desired hologram illumination.

Hologram designed for the modified illumination is used only to transform the spherical wave illuminating the hologram to the planar wave of the quiet-zone (QZ). The -1 dB beam width of the illumination defines directly the -1 dB beam width of the QZ, i.e., the DRFS beam defines the QZ size. The desired illumination has flat amplitude in the middle, corresponding to a 2 m QZ diameter, and -10 dB tapering on the hologram edge. Using the DRFS, the variation in slot widths in the hologram pattern is smaller compared to the previously used Gaussian illumination, which should improve the cross-polarization performance of the hologram.

The reflector surfaces were synthesized with a geometrical optics (GO) based synthesis method and optimized based on the simulations with physical optics (PO) and physical theory of diffraction (PTD). A DRFS has been previously designed with the same synthesis method for 310 GHz [4]. The designed 650 GHz DRFS has, despite the higher frequency, wider beam and significantly better beam quality. The simulations of the designed 650 GHz DRFS show excellent results. The -1 dB beam width diameter is about 2.34 m, which corresponds to a 1.96 m QZ diameter, and the hologram edge illumination is less than -10 dB. The maximum amplitude ripple in the -1 dB beam area is 0.45 dB peak-to-peak (0.24 dB rms) and the phase deviation is 5° peak-to-peak (0.8° rms).

[1] T. Hirvonen, J. Ala-Laurinaho, J. Tuovinen, A. V. Räisänen, A compact antenna test range based on a hologram, IEEE Transactions on Antennas and Propagation, Vol. 45, No. 8, pp. 1270-1276, Aug. 1997.

[2] A. Lönnqvist, T. Koskinen, J. Häkli, J. Säily, J. Ala-Laurinaho, J. Mallat, V. Viikari, J. Tuovinen, A. V. Räisänen, Hologram-based compact range for submillimeter-wave antenna testing, IEEE Transactions on Antennas and Propagation, Vol. 53, No. 10, pp. 3151-3159, Oct. 2005.

[3] J. Häkli, T. Koskinen, A. Lönnqvist, J. Säily, V. Viikari, J. Mallat, J. Ala-Laurinaho, J. Tuovinen, A. V. Räisänen, Testing of a 1.5-m reflector antenna at 322 GHz in a CATR based on a hologram, IEEE Transactions on Antennas and Propagation, Vol. 53, No. 10, pp. 3142-3150, Oct. 2005.

[4] J. Häkli, T. Koskinen, J. Ala-Laurinaho, A. V. Räisänen, Dual reflector feed system for hologram-based compact antenna test range, IEEE Transactions on Antennas and Propagation, Vol. 53, No. 12, pp. 3940-3948, Dec. 2005.

Paper describes design and optimalization of the septum polarizer in circular waveguide which will serve as a dish antenna feeder for the EME (Earth-Moon-Earth) station.

The feeder has two ports (one for transmitting and one for receiving), each produces circular polarization radiation with different sense (LHCP / RHCP), see Fig. 1 left.


Fig. 1.: The section-view of the 5-steps septum polarizer and its frequency response after optimalization

Requirements for such feeder are quite high: very low return loss, high isolation between ports and excellent radiation pattern polarization purity together with good symmetry of the pattern.
Because the structure is a bit complex to perform full-wave optimalization directly, we first optimized the radiation properties (mainly the polarization purity) of a 5-steps septum by the mode matching method. Then, using CST Microwave Studio, the whole structure was simulated and coaxial-to-waveguide transition has been optimized to match at 1.296GHz (see Fig. 1 right). Simulated results are very promising, isolation between ports is S21=-26.3dB @ 1.296GHz and Return Loss S11= -47.8dB @ 1.296GHz. Radiation pattern exhibit great rotational symmetry and LHCP / RHCP patterns have 40dB spacing at the main lobe (see Fig. 2).

Fig. 2.: Simulated radiation pattern – LHCP / RHCP cuts (0,45,90,135°) when the LHCP port is excited

In full paper we will show measured results (which fits very well with simulated ones) also and several calculations of the polarization efficiency for the feeder. We are currently optimizing chokes which will form the radiation pattern to help to illuminate the dish better together with high polarization purity.

State of the art:

Present and next generation telecommunication satellite systems often require multiple beam capability. The three fundamental ways to achieve high coverage gain are overlapping feed arrays, with complicated feed networks, interleaved beams, with multiple apertures, and feeds with high aperture efficiency, where spillover loss is traded against beam roll-off. In the last case, which is addressed here, we need a small feed element with high aperture efficiency, i.e. high directivity and narrow beamwidth. G. V.

Trentini [1] proposed to use a partially reflecting screen in order to increase the feed directivity. A point source would generate an ensemble of waves that impinge on a screen that is partially reflective and partially transmitting. The reflected rays propagate laterally and then are redirected toward the partially reflecting screen, so that overall they generate a unique leaky wave that mainly radiates in the broad side direction. More recently [2] the same concept has been applied to multi-beam reflector antennas. A similar effect can be obtained by using dielectric super-layers [3]-[5]. As is the case with most leaky wave configurations, these attempts are characterized by narrow frequency bandwidths.

This contribution

deals with the same problem, i.e. the design of dielectric super-strates to enhanced performances of multi-beam reflector antennas. In this scope the leaky poles of the Green’s function are studied and their impact on the bandwidth and mutual coupling. It is found that both the input and mutual admittance are not fast varying functions of the frequency when the leaky wave beams are pointing away from broadside. This allows a different trade-off between the directivity enhancement and the applicable frequency range. An array of 5x5 elements has been designed and will be manufactured showing a predicted improvement, with respect to the free space case, of 1.6dB over a 6% bandwidth in the Edge of Coverage gain (which is defined as the cross-over gain between three adjacent beams).

References:

[1] Trentini, G.V., "Partially reflecting sheet arrays", IEEE Trans. AP, vol.4, no.4, Oct 1956, pp. 666- 671.

[2] S.I. Khurein Castiglioni, G. Toso, C. Mangenot, "Multi-beam Antenna Based on a Single Aperture Using Overlapped Feeds", Proceedings of 13-th JINA November 2004.

[3] D.R. Jackson, A. A. Oliner, Antonio IP "Leaky Wave propagation and radiation for a Narrow-Beam Multiple layer Dielectric Structure" IEEE Trans. AP, Vol.41, no.3 March 1993, pp. 344-348.

[4] C. Cheype, C. Serier, M. Thevenot, T. Monediere, A. reineix, B. Jecko, " An Electromagnetic Bandgap Resonator Antenna" IEEE Trans. AP Vol.50, no.9 Sept. 2002, pp. 1285-1290.

[5] Guerin, N.; Enoch, S.; Tayeb, G.; Sabouroux, P.; Vincent, P.; Legay, H. "A Metallic Fabry–Perot Directive Antenna" IEEE Trans. AP, Vol. 54,no. 1, Jan. 2006 pp. 220 - 224.

In this work, the two- and multi-level Fast Physical Optics (FPO) algorithms recently proposed by the authors (A. Boag and C. Letrou, IEEE Trans. Antennas Propagat., 51(5), 1063-1068, 2003 and 53(6), 2064-2072, 2005, respectively) are combined to compute the radiation patterns of dual reflector antennas.

In these fast algorithms, the radiating surface is decomposed into subdomains whose radiation patterns are fully described by their samples computed on a coarse grid of directions. Interpolation and aggregation of these patterns are then performed in the relevant coordinate systems, to obtain the fields radiated by the whole radiating surface on a fine grid of directions, either at once (in the two-level algorithm) or after recursion through successive levels (in the multilevel algorithm).

FPO algorithms have been shown to reduce the complexity of Physical Optics (PO) integral computations, respectively, to O(N³) (two-level algorithm) and to O(N²log N) (multilevel algorithm) instead of the original O(N⁴), for a problem of electrical size N=kR (k is the wavenumber, and R is the radius of the smallest sphere circumscribing the radiating surface). Moreover, the integrations can be effected very accurately, independently of the size of the problem: they are performed on subdomains whose size can be taken as small as necessary for the chosen integration routine to reach the prescribed accuracy. The accuracy is then directly related to the errors induced by the interpolation process.

Assuming that the radiation pattern of the feed is known, we shall focus on the two following steps in the conventional PO analysis of a dual reflector antenna:
(a) computation of the fields and surface currents on the surface of the main reflector due to the radiation of the sub-reflector, illuminated by the feed;
(b) evaluation of the far fields radiated by these surface currents present on the main reflector.
Regarding the accuracy of computations, the case of dual reflector antennas is considerably more challenging for FPO algorithms than the case of single reflector antennas for two reasons.
(i) In the first stage, interpolation-aggregation must be effected at finite distances from the radiating surface of the sub-reflector, and no longer in the far field approximation. This computation is be performed by the two-level FPO algorithm.
(ii) Computation of the final radiation pattern is performed through interpolation-aggregation (multilevel algorithm) of fields radiated by current sources pre-computed on the main reflector through the interpolation-aggregation process performed at step (a). The algorithm thus involves two nested interpolation-aggregation steps.

The performance of the resulting algorithm will be demonstrated on typical offset dual reflector designs and its accuracy will be discussed.

Fresnel Zone Plates (FZP) substitute the curved surfaces of a Fresnel Reflector for an stepped one, getting advantages concerning to manufacturing and prize. Besides, this technology provides the possibility of designing conformal antennas specially in case a double beam is required. The main disadvantage of FZP is the inherent loss in gain of the stepped approximation, though, as we have been able to prove, efficiency results to be very reasonable and equilibrated with the other advantages for many uses.

Conic Fresnel Zone Plates (CFZP) consist of FZPs developed over a conic surface instead of a planar one. This change provides more efficiency preserving the advantages of FZPs without a noticeable increase in size or occupancy.

The purpose of this paper is to present a response analysis of CFZPs in comparison with same aperture FZP, extracting their behaviour as a function of the parameters of the CFZP. Analysis will be done making use of an efficient simulator developed by the authors with the purpose of evaluating ring-decomposable reflectors, as is the case.

At the same time, a procedure for optimal synthesis of CFZPs will be presented.

Jodrell Bank operates and array of large radio telescopes, separated by up to 200 km, called MERLIN, which allows astronomers to produce radio maps of the sky with resolution equivalent to that of the Hubble Space Telescope at optical wavelengths. Astronomical observations are carried out in a variety of frequency bands and, in order to maximize observing efficiency, it is desirable to be able to switch remotely between frequency bands. As part of this frequency flexibility there was a need to develop a remotely deployable feed system, covering the frequency range 1.33 to 1.73 GHz, to operate at the secondary focus of a 25 metre diameter cassegrain reflector antenna. The chosen solution was a relatively small corrugated conical horn in combination with a polyethylene lens approximately 1.5 metres in diameter. A turret arrangement in the antenna's vertex cabin allows the horn to rotate into position as required, or to be replaced by higher frequency feed horns. A separate mechanism moves the lens into position when required, and stows it out of the way when the antenna is used at higher frequencies. This paper describes the optical design of the lens and horn and the physical realization of the feed system. Details are also given of tests carried out on scale model feed systems and on the full size feed. Three antennas will be equipped with this feed design; to date one antenna has been fitted with a feed system and performance results will be presented. Overall aperture efficiency was over 50 %, meeting the target and exceeding the performance of a rival horn-only design for a similar application. Some problems were encountered due to the pick up of thermal noise inside the vertex cabin, but these are being resolved by additional screening.

We introduce a partly corrugated hard horn, consisting of a smooth-walled horn with an attached longitudinally corrugated outer section. This alleviates the problems with the manufacturing when the longitudinal corrugations extend into the throat of the horn. The transition between the inner smooth walled part and the outer corrugated part is abrupt. This is used and controlled to design better and shorter single-band horns than otherwise possible. Furthermore, it enables the design of dual-band horns with low cross-polarization and high gain, for multi function use at Ka-band with transmit and receive frequencies in the same antenna. A model of such a dual-band horn was manufactured at Saab Ericsson Space and measured.

The design has been done by using the efficient mode matching software that has been developed by Skobelev in collaboration with Chalmers (based on using a homogenized anisotropic wall model). This does not take the finite period of the corrugations into account, but will predict the radiation characteristics correctly if the period is small enough. This has been verified by heavy computations using a 3-D FDTD code. The measured laboratory model had too large cross polar sidelobes, but the later extensive computations showed that these sidelobes would have been in agreement with the mode matching results if the number of corrugations were increased.

The project has been funded in part by ESTEC in connection with a project on multimedia satellite systems at 20 and 30 GHz.

[1] O. Sotoudeh, P.-S. Kildal, P. Ingvarson, and S. P. Skobelev, "Single- and dual- band multimode hard horn antennas with partly corrugated walls", accepted for publication in special issue on multifunction antennas in IEEE Transactions on Antennas and Propagation, Feb 2005.

A conventional GPS (Global Positioning System) receiver antenna would have the RHCP (right hand circular polarization), an upward pointing radiation pattern, high gain of about 4 dB and axial ratio of 3 dB max. Back lobes are undesired. The centre operating frequency is 1575.42 MHz and the expected bandwidth is10 MHz and. This kind of GPS receiver antenna with C/A codes works well in the outdoor environment. However, the problems, such as signal failure or less accuracy, were posed when it goes to the urban or indoor environments because of the multipath fading [1]. The current solution for indoor-capability GPS, or more precisely high-sensitivity GPS, is a combination of Assisted-GPS (A-GPS) and massive parallel correlation [2]. Although wireless networks can potentially provide timing and ephemeris assistance to an embedded GPS receiver, some fundamental business, technical, and performance challenges continue to drive GPS manufacturers' desire to not only minimize their dependence on network aiding, but also improve location reporting performance and robustness.

In this paper, a 2-element pattern diversity antenna array is proposed for the Galileo receiver to offer better multipath performance. It is modelled by using the CST Microwave Studio^TM package, which utilises the Finite Integral Technique for electromagnetic computation. The proposed design of the diversity antenna array consists of two helical antenna operating at 1575.5 MHz. Each element is made by winding 0.5 mm diameter copper wire around diameter gauge of 2.5 mm diameter with a height of 15 mm. It has also been found that the polarization requirement of RHCP is desired, but may not be a strict requirement due to highly scattered environment [3]. Therefore, the linear polarization is selected for the proposed diversity antenna array. The proposed pattern diversity antenna array is designed to receive the satellite signal from different directions by tuning the radiation pattern of each element. The diversity performance, such as correlation coefficient, mean effective gain and diversity gain, of the proposed antenna array is also studied. It is found that the proposed 2-element pattern diversity antenna array can obtain 8.43 dB diversity gain than the single helical antenna case. This can offer 8.43 dB link margin to the Galileo system.

References:

[1] G. Dedes, A.G.Dempster, "Indoor GPS positioning - challenges and opportunities", Vehicular Technology Conference, VTC-2005-Fall. 2005 IEEE 62nd Volume 1, pp: 412-415, 28-25 Sept., 2005.
[2] F. van Diggelen, "Indoor GPS theory & implementation", Position Location and Navigation Symposium, 2002 IEEE, pp: 240-247, 15-18 April 2002.
[3] V. Pathak, S. Thornwall, M. Krier, S. Rowson, G. Poilasne, L. Desclos, "Mobile handset system performance comparison of a linearly polarized GPS internal antenna with a circularly polarized antenna", IEEE Antennas and Propagation Society International Symposium vol.3, 666, 2003.

Two circularly-polarised wires antennas, operated at 1600 MHz and intended for satellite communications applications, were designed using Genetic Algorithms (GAs). These are the Quadrifilar Helical (QHA) and Quadrifilar Spiral (QSA) Antennas. The antennas are firstly considered as a test on an infinite ground plane and then optimised further on small size handsets. The attained optimal antenna geometries for each of the antennas are presented in Figs. 1 and 2. The capabilities of GA are shown as an efficient optimisation tool for selecting optimal parameters to be used in simulations with an electromagnetic antenna design code, seeking convergence to designated specifications. The performance of the optimised antenna designs was analysed using a commercial simulator in terms of VSWR, axial ratio and power gain, as sample of these results shown in Figs 3, 4, 5 and 6. Presented results indicate that the optimal antennas met design objectives under several certain constraints. A further study on the achievable of a minimum size of these antennas to build on a handset is in progress. This also includes the axial ratio purity of these antennas when they are operated next to human head. Performance of helical and spiral antennas with two and three arms is investigated and discussed in the presence of mobile handsets.

The sequential switching of elements in antenna arrays in, e.g., angle-of-arrival (AoA) estimation systems is an attractive option to eliminate the requirement for multiple receivers. The undesirable mutual coupling between the elements in the array, however, remains present. Fig. 1 shows a circular switched antenna array geometry used in an AoA estimation system.

In AoA estimation systems, mutual coupling (MC) effects can be avoided with the aid of virtual arrays, in which a single antenna is moved in space, for example along a linear or circular trajectory. This technique can, however, only be used for stationary measurements and requires advanced mechanics to create array geometries other than linear or circular.

In switched antenna arrays the coupling of the active antenna with the passive antennas can be reduced by minimizing the radiated power of the passive antennas. To minimize the re-radiated field in all directions, we propose a method to determine an impedance Z_L for a comparable idealized monopole that minimizes the induced current magnitude by changing the electrical dimensions of the antennas through the impedance in which their feedpoints are terminated.

In order to verify the theory, simulations are performed on a two-element antenna array positioned in the horizontal plane using different terminations on the passive element. The antenna elements consist of vertically polarized drooping radial monopole antennas and are designed to be resonant at 2250MHz. The antenna input reflection parameter S₁₁ and δ, the maximum distortion in the antenna pattern, of the active antenna are analyzed for different complex terminations of the passive antenna. Using the theoretical method, the value of the termination is determined to be Z_L = j245Ω. Simulation results show that the lowest values for S₁₁ and δ are obtained by terminating the passive antenna in an impedance close to j250Ω.

Measurements were performed in an anechoic chamber with two antenna elements in a configuration identical to that employed in the simulations. It is shown in Fig. 2 that the distortion in the antenna radiation pattern can be reduced from about 6 dB to less than 2 dB if the passive antenna is terminated using a tuned stub that represents the reactive load, instead of 50Ω. It has also been verified that this technique is effective when a combination of two switches is used to alternately load or select elements as would be required in an actual array implementation.

From simulations it was found that the optimum impedance value depends mainly on the geometry of the antenna elements, and is quite independent of the distance between the active and passive antennas. This makes the technique suitable for switched antenna arrays consisting of more than two elements. The technique is applied to the switched antenna array in Fig. 1 and can readily be applied to switched arrays of linear antennas in general. This paper will present a novel method to reduce the mutual coupling effects in switched antenna arrays by changing the feedpoint termination. Apart from the theory, a practical method to obtain the optimal impedance is suggested supported by simulations and measurements.

Satellite uplink antennas for SHF band communications may require high resolution adaptive nulling capability to provide sufficient pattern gain to desired users while mantainig pattern nulls on interference sources in close proximity to the users.

Multiple beams antennas (MBA) consisting of an aperture illuminated by a collection of feeds located in its focal plane and a beam-forming network (BFN) for combining the outputs of the feed array are well-suited as nulling antennas for geosynchronous satellites.

Demanding nulling performances in terms of interfering signals cancellation, require an accurate pattern prediction down to sidelobe level and an high cross-polar isolation of the radiated element beams. To meet these requirements, design methods and analysis tools shall be employed that takes into account the electromagnetic interaction among the feeders and the satellite structures.

In the qualification phase of an SHF nulling MBA at AASI, the pattern measurements have shown an unpredicted cross-polar level due to the interaction of primary radiated field with the surrounding structures.
This paper reports the main steps of the AASI and IDS activity in optimizing the antenna position and the shape of some antenna-farm structures in order to improve the cross-polar performance taking into account the mechanical constraints. The electromagnetic analyses have been carried out employing different numerical techniques in order to validate the results and reduce the risks and the cost of satellite structural modifications. The numerical techniques, the optimization procedure and measured results are reported and discussed.

Interaction between electromagnetic sources closely coupled to highly lossy media has been studied extensively in recent years, especially in the areas of cellular communication, medical devices, and hyperthermia-based treatments (e.g., R. W. P. King, J. Appl. Phys., 87, 893-900, 2000; G. Lazzi et all., IEEE Trans. MTT, 52, 1853-1855, 2004; A. J. Fenn et all., Int. J. Hyperthermia, 15, 45-61, 1999). Most studies consider specific configurations with variable geometric and physical parameters without making clear separation between contributions of various polarizations generated by a radiating element. In general, an arbitrary source orientation can be expressed in terms of a finite number of fundamental polarizations, each characterized by inherently different electromagnetic characteristics, particularly those associated with local and total Specific Absorption Rate (SAR) behavior.

Herein, we focus on the analysis of electric and magnetic dipoles interaction with semi-infinite highly lossy biological medium, studying both vertical and horizontal polarizations. This class of canonical models has been recently shown as an effective mean for obtaining physical insight into the basic power absorption mechanisms as well as tight bounds and estimates on power relations and SAR (D. Razansky et all., IEEE Trans. EMC, 47, 2005; D. Razansky et all., J. Appl. Phys., 95, 8298-8308, 2004; D. F. Soldea et all., J. Appl. Phys., 94, 2053-2059, 2003). The analytic study is carried out by decomposing each polarization configuration into transverse electric and transverse magnetic modes (L. B. Felsen and N. Marcuvitz, Radiation and Scattering of Waves, IEEE Press, 1994), resulting in closed-form and readily interpretable expressions for basic power relations, a distinct feature of our approach. In particular, these results are shown to be strongly dependent on the evanescent spectral content of each mode. The vertical magnetic dipole polarization contribution to the total SAR is thus expected to dominate due to its largest evanescent spectrum content. Further implementation of this modeling approach enables a better understanding and an effective design of practical systems comprised of sources radiating in the vicinity of living tissues.

The analysis of the mutual coupling effects in antenna arrays by solving Maxwell's equations numerically requires large computational time and resources, and hence is only suited for arrays of moderate sizes. To reduce the time required to calculate the mutual admittance between apertures, interpolation can be used. In interpolation, the mutual admittance between a small number of aperture pair configurations is calculated, and the results are used to determine the mutual admittance between any aperture pair configuration in the plane.

In the proposed approach, the mutual admittance between two apertures in a common, flat, perfectly conducting ground plane is calculated with an improved version of the equations proposed in [1]. Nine element pair combinations are required, whose coupling admittances and relative positions are known. The admittance values can be obtained, by either calculations or measurements, to establish a set of nine independent equations that are then solved to obtain nine complex interpolation coefficients. The equations can then be employed to determine the mutual admittance between any aperture pair configuration in the plane.

The modelling approach is implemented using a computer program in Matlab. It can employ both numerically evaluated and measured data. For illustrating the capabilities of this approach, the mutual couplings in a 63 element shared aperture antenna (see Fig. 1) were evaluated. Initially, the method's robustness was assessed by means of comparisons with the full set of numerically computed mutual couplings. The values of the mutual admittance between a reference element and nine test apertures were used to form nine complex linear equations that were then solved for determining the required complex interpolation coefficients. Of the nine apertures, three are placed in the H-plane, three in the E-plane and three between the two planes. Finally, the interpolated values were compared against the original data and the observed deviations did not exceed 0.5% for the dominant (TE₁₀) mode.

After gaining confidence in the method's robustness, the strategy was applied to the case of physically measured couplings. To this end, the couplings between 1_L and all other elements in the L sub-array (the grey one in Fig. 1) were measured. The comparison between the measured and interpolated results is depicted in Fig. 2. From it, it can be easily concluded that the interpolation strategy is highly accurate.

References

[1] M.C. Bailey, "Technique for Extension of Small Antenna Array Mutual-Coupling Data to Larger Antenna Arrays," NASA Technical Paper 3606, August 1996.

The work presented in this paper is the analysis of out of band coupling to aircrafts antennas. Several technologies and operational bandwidths have been considered. The assessment technique, combines measurements and calculation techniques. The technologies considered include blades, monopoles, patches and loops antennas in the VHF, L and C bands. Most of these antennas radiate an omni directional pattern. Typical locations were chosen on top and bottom of the aircraft shown on figure below.

The information required to assess the coupling are the antennas s parameters and their patterns. The s parameters were measured in the laboratory on a large bandwidth. As regards the s11 parameter, the antennas are found to operate at two bandwidths: around the nominal frequency and around the third harmonic frequency where depending on the technology, the value is usually still acceptable.

The antennas patterns were then either measured for the first three harmonic frequencies or numerically computed depending on the capability to perform the measurements. In the last case, the calculations were carried out using the fast multipoles technique (up to 1 GHz) and the UTD asymptotic technique for higher frequencies.

The results obtained for the coupling of the different antennas to external fields will be presented.

VHF and UHF antennas installed aboard small and medium size satellites usually strongly interact with the bus structure and with appendages, due to the fact that their dimensions can be resonant with respect to the antenna wavelength. The performance of the antenna can be therefore made significantly worse and requirements can be also no more satisfied due to the electromagnetic interaction among the radiator and the superstructures. Moreover, Tx antennas which radiate high power can cause RS (Radiated Susceptibility) problems on equipment (e.g. IRES) located on the satellite top-floor, due to the fact that such equipment are electrically near to the antenna and to the resonant appendages.

Alcatel Alenia Space and IDS managed a problem in which a UHF array was designed and then installed aboard a TLC satellite. Each one of the aforementioned problems at installation level has been analyzed and recognized by means of numerical modeling of the whole array and satellite.
Each problem has been solved by optimizing the array position and by means of a "fine" tuning of the array orientation.

In this paper a description of the RFC problems related to V/UHF radiators installed aboard spacecrafts is reported, focused on the before mentioned real life application. The numerical models and the applied optimization procedure are reported and discussed.

The mutual coupling between adaptive antenna array elements degrades the performance of the array especially for direction finding purpose. There are many possibile ways to compensate the coupling effects via electromagnetic analysis or via signal processing. We are following the first approach, calculating the mutual coupling matrix and compensate its effects. The mutual coupling is investigated for microstrip structures. The investigation gives important results, because the different coupling in x and y direction in the planar array gives an explanation on the asymmetric radiation pattern. Based on the results also the compensation of the mutual coupling between array elements can be corrected for minimising the DoA estimation error. Introduction The simulation for the arrays especially for the 4 by 4 array is based on the element by element analysis because of the large structures. Our microstrip antenna array has an asymmetry in x and y direction in excitation as can be seen in Fig. 1.

The microstrip structure is analysed using Ansoft Designer and with two programs, a FEM-MoM program and a FDTD program developed at our department. The FEM solves the weak form of the three dimensional integral equation using tetrahedron test functions and again Galerkin approach is used. On the border of volume analysed a PML boundary condition is used. Fig. 1. The patch geometry investigated Fig. 2. Mutual coupling (S12)between patch elements and input reflection for two positions in array

Results

The analysis was performed for two antenna pairs in array (Fig. 1.) The calculations shows a difference in mutual coupling in the two directions namely there is a bigger coupling between elements in a direction and this effect also causes an increasing input reflection. In b direction because the decreasing coupling the input reflection is also much better. Based on this calculation it can be explained the difference in radiation pattern in the two planes. The presentation shows comparison for the three analysis methods and with our measurements.

Reference

D. Segovia-Vargas, R. Martin-Cuerdo, M. Sierra-Perez: Mutual coupling effects correction in microstrip arrays for direction-of-arrival estimation, IEE Proc. Microw. Antennas Prop., Vol. 149, No 2. pp. 113-118, 2002

Patch and slot radiation with grounded dielectric slab is well studded. Surface waves excited with grounded dielectric slab cause decreasing of space wave radiation efficiency of patch and slot antennas. Moreover surface waves increase interaction between antenna array elements. Fabrication and experimental verification of materials with negative value of permittivity and/or permeability shows interest in patch antenna design with metamaterials. Usually TE and TM wave expansion of layered structures is used for calculations of patch and slot impedance and radiation. The poles of integrand are associated with surface waves. These poles complicate calculations of antenna impedance and space wave radiation. Using alternative wave expansion with LE, LM waves it is possible to separate space and sur-face waves contributions. There are no any poles in integrands. This fact simplifies the problem. The Green's functions of grounded magnetodielectric slab for this case are suggested. Space wave radiation efficiency and surface wave radiation patterns are calculated for dipole on metamaterial grounded slab and slot covered with metamaterial. Radial propagation constants for electric and magnetic surface waves are determined. It was very interesting to compare surface wave properties for ordinary dielectric substrates and substrates with metamaterials. There is always at least one sur-face wave in ordinary substrates because it does not have any cutoff frequency. A number of sur-face waves depend of dielectric permeability and thickness but the first wave is LM type. In meta-material substrates the order excited surface waves is changed. In some situations the first wave is LE type. Under certain conditions there is no surface wave at all. The radiation pattern difference between LE and LM surface waves in ordinary substrates and substrates with metamaterials is ana-lyzed. Directions of high interconnection between antenna array elements are predicted. Conditions of high efficiency of space wave radiation of patches and slots are defined.

In this paper, a study is presented whose goal is to develop an evaluation method of the human exposure to electromagnetic fields in terms of the specific absorption rate (SAR). The challenge of this study is the direct evaluation of the SAR through on site measurements of the incident electric fields. Straight on site SAR evaluation is very difficult, thus a hybrid technique such as that proposed in [1], combining in-situ measured incident fields and FDTD [2] calculation of the SAR, is more suitable. For this purpose, after reviewing some constraints, we adopt the plane wave spectrum as an efficient mean to couple the measurement to the FDTD calculations via the Huygens box. A planar scanning is more pratical especially on site. The plane wave representation allows evaluating the electric and magnetic fields at any distance (in the near or far field zone) and especially on the Huygens box that contain the phantom inside which the SAR would be estimated (Figure1). This approach has the advantage of being more realistic as the source and the environment are characterized by means of measurements. Furthermore, FDTD solutions for the normalized incident plane waves can be pre-calculated. As the plane wave spectrum is determined by measurements, no more FDTD simulations would be necessary to evaluate the SAR. We further study the relation between the incident plane waves and the induced SAR in order to reduce the spectrum required for the main contributions.

Measurements have also their limitations (phase measurements difficulties, fading, uncertainty ...). So methods are developed in order to be able to exploit optimally measurement data. In particular, phase measurements are very challenging on site and only intensities can be measured. Therefore, phase-retrieval-based approaches are considered involving the solution of a non-linear inverse problem.

[1] Om P. Gandhi: "An On-site Dosimetry System for safety assessment of wireless base stations using spatial harmonic component" IEEE Om P. Trans. Ant. Prop. , vol. 51, no 4, pp. 840-847, April 2003.

[2] A.Taflove (1995, June). "Finite-difference time domain (FDTD) techniques and applications in computational electrodynamics".

A topic of significant interest is how and at what levels energy from mobile phones may interact with the head and in particular the eyes. Recently personal data assistants (PDAs), with integrated mobile phones have begun to be popular. Unlike previous devices these PDAs are held in front of the face and recently the authors have discovered that significant interaction can occur between RF energy of the type generated by mobiles and jewellery, for example spectacles. This paper investigates the effects of metallic jewellery on the Specific Absorption Rates (SAR) in different models of the human head. Metallic jewellery to increase SAR in the head however, the levels of interaction depends on the shape and the electrical properties of the tissues in the head. A metallic pin was found to increase the SAR averaged over 1g by 13.5 times in a homogeneous cubic head, whereas, the same pin increased the 1g SAR in an anatomically realistic head by factor of 2. A half-wavelength dipole excitation at 1800MHz is positioned in front of the head to represent a cellular enabled personal communication device. The results shown in this paper are for a metallic pin. This could represent facial piercings, a section of metallic spectacles or a microphone for a hands free kit. In previous research, the authors [1] have found that metallic spectacles can significantly increase the SAR in the eye. An independent 3D FDTD code [1] was used to model the metallic jewellery in proximity to the human head. The DASY4 measurement System was also used to verify the results. Figure 1 shows results of the SAR along an axis perpendicular to the pin, and 3mm inside a homogeneous cubic phantom with a 2mm fibre glass shell, using both the FDTD code and the DASY4 System. The measurements provide good agreement with the simulated results. The figure shows that the pin causes significant increases to the SAR at the pin location but the presence of the pin decreases the SAR towards the edges of the cubic head. Results from the FDTD code showed that the effects of the jewellery are dependent on the electrical properties and geometry of the head. Figure 2 shows results of a horizontally orientated dipole at 1.8GHz positioned in front of a realistically shaped head. A 70mm horizontal pin added by the forehead increases the 10g SAR by 2 in an anatomically realistic head but the same pin increases the 10g SAR by 3.5times in a homogeneous version of the same shaped head. Figure 1. A 70mm dipole 98mm from a cubic phantom with a 70mm metallic pin. SAR is calculated 3mm into the homogeneous brain for the FDTD simulations and measured 3.6mm into the liquid for DASY4 results. Figure 2. The relative enhancement in the 10g SAR (with pin/no pin) caused by different lengths of horizontal pin situated by the forehead of different head models. Note the external geometry is identical in all 3 cases.

References

1.Whittow, W.G. and R.M. Edwards, A study of changes to specific absorption rates in the human eye close to perfectly conducting spectacles within the radio frequency range 1.5 to 3.0GHz. IEEE Trans. Antennas and Propagation, 2004. 52(12): p. 3207-3212.

Microstrip patch antennas are suitable for medical applications due to light weight, compactness and conformability [1]. Antenna miniaturisations plays a vital role in the design of hyperthermia applicators for the treatment of small tumors situated on curved sites of the body surface. A compact circular patch antenna embedded in a narrow annular-ring is examined in terms of return loss, radiation efficiency and SAR when in proximity to human tissue. The antenna employs a crossed slot in the ground plane. This recently reported novel geometry [2] yields a significantly smaller size for a given frequency compared to conventional square or circular patches. The size reduction is about 50%. A further augmentation of this antenna was made for hyperthermia applications by adding an annular slot in the ground plane, which increases the power deposition in the body by 0.5 dB. It can also be placed directly on the body without significant detuning, which occurs with conventional microstrip geometries, without slotted ground plane. In the study, the proposed antenna is compared with the conventional circular patch antenna. Both antennas were placed at distances between 0 mm to 50 mm from the human tissue, which comprised transversal muscle fiber. At 434 MHz, the muscle properties are 56.86, 0.805S/m and ñ = 1040kg/m3. The muscle slab dimensions are 300mm x 300mm and 100mm thick.

Fig. 1 Geometry of proposed patch antenna
a Profile, b Ground plane, c Patch on top

Fig. 2 S11 for conventional and proposed patch in free space and in proximity to tissue

The presence of muscle tissue at the patch surface decreased the return loss in the conventional circular patch antenna from 22dB to 4dB at the operating frequency, while the return loss remains relatively constant for the proposed antenna. This is seen in Fig. 2. The SAR is therefore greater for the proposed antenna, as it can be placed closer to the tissue. A full evaluation of the proposed antenna will include internal field components, efficiency and SAR distribution.

References

[1] F. Montecchia, Microstrip-Antenna Design for Hyperthermia Treatment of Superficial Tumors, IEEE Transactions on Biomedical Engineering, 1992, 39 (6), 580-588.

[2] X.L., Bao, M. J.,Ammann, Compact annular-ring embedded circular patch antenna with cross-slot ground plane for circular polarization, Electronics Letters, 42, (4), 2006, 192 – 193.

We have developed a clinical microwave imaging system for breast imaging applications including cancer detection and monitoring of neoadjuvant chemotherapy. For the latter, key features of any montoring system must include the ability to image a target in a repeatable fashion while being noninvasive and relatively inexpensive. Our microwave imaging system is particularly useful in this setting because we have overcome challenges such as the use of a priori information with the combination of our unique imaging configuration and software. The hardware incorporates monopole antennas which present isotropically radiated fields within which complex nulls (and accompanying non-unique signal phase distributions) are generally not generated until the electromagnetic wave has propagated beyond the illumination zone and associated receiver antennas. This is particularly useful for our variance stabilizing transformation algorithm which can recover accurate dielectric property maps without a priori information or converging to local minima as long as the unwrapped phase distribution remains unique. The associated unwrapping of the measurement data can be readily performed through the use of multi-frequency data and standard unwrapping algorithms.

In this paper, we describe our clinical system and the first set of images from a patient undergoing chemotherapy. She received four doses of doxorubicin and cyclophosphamide once every three weeks and was imaged with the microwave system during her first three hospital visits. Subsequently she also received a second round of chemotherapy once every two weeks for six weeks after which we were able to image her a last time. She was also imaged with gadolinium-enhanced MR at the start of the chemotherapy and just prior to surgery. The patient was 36 years old, her breasts were considered radiographically scattered density and the tumor was estimated 6.5 to 7.0 cm in diameter prior to the start of the treatment. The microwave images clearly show a distinct reduction in the elevated property (both permittivity and conductivity) zones associated with the tumor location. These were consistent with the clinical assessment where the oncologists concluded that the tumor had responded completely to the treatment based on the MR images and on conventional CT images examining the size of the axillary adenopathy.

This is an exciting result in that it demonstrates our ability to image a large, high contrast object within an in vivo, low permittivity organ and recover reliable and repeatable images. This may become an important clinical application for microwave imaging given the limited tools currently available to breast oncologists.

The need in mobility and in higher data rate transmission result in the shift of operating frequency of new wireless communication systems towards the millimeter wave (MMW) frequency band. Frequencies around 60 GHz attract particular attention for broadband indoor short-range communications. Recently, these frequencies have been clearly identified as highly promising for next generation WLAN and WPAN systems. In this context, special attention should be payed to potential biological effects of 60 GHz exposures artificially induced in our environment by new wireless communication systems.

The objective of this study is to investigate the influence of low-power MMW radiation at 60 GHz on the structural state of phospholipid layers in the biological membranes. According to the concept of modern biophysics the most realistic membrane model is represented by double lipid layer where the proteins are integrated. Phospholipids are the most prominent in quantity lipid constituents of biological membranes. Artificial phospholipid membranes were exposed or sham-exposed at 60 GHz for various durations and the structural state of the phospholipid monolayer was characterized by the measurements of superficial pressure dymamics of phospholipid monolayers.

To ensure a sufficiently homogeneous field distribution at the membrane level an exposure system based on both pyramidal and conical horn antennas has been developed. The field distribution at the membrane surface has been calculated theoretically and compared with measurements. The role of various radiation parameters, namely incident power density, polarization, amplitude modulation, different time regimes of exposure, have been investigated.

The main outcomes of our work are the followings. An increase of superficial pressure of phospholipid monolayers was evidenced for diverse experimental configurations. The data obtained for different superficial power densities have not revealed any particular role of the exposure parameters, such as polarization, amplitude modulation or time-interrupted regime of exposure. However, experimental results have shown that even very low power densities of MMW radiation at 60 GHz lead to significant lateral pressure increase. To explain observed effect we have focused our attention on the two hydrocarbon moleculear groups largely represented in biomembranes. These groups are of the particular interest because of their rotational spectral lines situated close to 60 GHz: (i) HCCCCH with the spectral lines at 59.677 GHz, 59.846 GHz, 59.937 GHz (Matsumura et al., J. Mol. Spectrosc. 96, 219, 1982); (ii) CH₃CH₂CCH with the spectral lines at 60.047 GHz, 60.151 GHz, 60.379 GHz, 60.490 GHz, 60.498 GHz, 60.734 GHz, 60.762 GHz (Landsberg et al., J. Mol. Spectrosc. 98, 210, 1983). We suppose that changes in superficial pressure dynamics of phospholipid monolayers after exposure to 60 GHz are due to the excitation of hydrocarbon groups in apolar part of the phospholipid molecules.

I. Introduction

In electromagnetic dosimetry mobile phone antennas is typical problem and consequently far-field exposure receives less attention. International safety limits provide reference levels expressed in terms of EM field strengths, which are evaluated in the absence of a person and derived from plane-wave incidence and CW exposure, which is limited to far-field situations. The rationale for deriving basic restrictions and their associates safety margins is not fully standardised and diverse values are employed depending upon the thermal effect being considered. In this contribution more realistic exposure scenarios and response effects are evaluated, including a 10W/cm2 plane-wave incident exposure. SAR is provided for all tissues.

II. Methods and models

In order to be able to measure SAR values, homogeneous models are required wherein a solo tissue is characterised by averaged properties over a certain number of tissues. Heterogeneous models of man have been used by selecting several tissues. While for near-field exposure first ones represent an overestimation of SAR respect to that obtained with second ones, for far-field plane-wave exposure the situation depends strongly on the frequency of operation and the portion of the body under test. For the human head, considerable higher both electric field and SAR values are obtained with second ones. The head model we have employed represents the tissues found in the Visible Human Project directly from the ear reference point (EPR) towards the inside, as depicted in figure 1. Homogeneous equivalent models have also been employed for comparison purposes.

Fig. 1. Selection of heterogeneous model tissues.

Current safety limits are derived for averaged SAR under near-field exposure, but in order to assess possible health hazards, thermal safety factors have been proposed by computing temperature increases in the model and comparing directly to adverse thermal effects. Thus, the human thermoregulatory response to RF energy absorption has received attention recently. In this contribution Maxwell equations have been hybridized to heat and mass transfer equations through a modified bioheat equation:

The thermal model includes heat diffusion and convection, metabolic heat production and heat-sink from tissue volume by blood perfusion, so that thermoregulatory control is achieved in the model, which can keep a constant temperature under no RF exposure.

III. Simulated results

As expected, the heterogeneous model provided both higher peak SAR and Electric field deposition, particularly in the skull, but temperature increase was smaller in the cerebellum area respect to various homogeneous models, illustrated in figure 2 and listed in Table 2. Thus suggests that the real multilayer tissue structure for human beings, particularly the location of the skull, acts as a thermal protector for the inner cerebellum and brain respect to incident electromagnetic fields.

Fig. 2. Electric field distribution and temperature increase in the simulated models.

Table2. Simulated results.

IV. Conclusions

Hybridisation between EM field exposure and thermoregulatory response has provided interesting results, such as the additional thermal safety factor that skull represents for human body under EM field exposure. The described model could be useful for surgery operations like BDS, etc.

The initial step in photosynthesis is the perception of light (an electromagnetic radiation) by plant leaves. Plant can therefore be assimilated to electromagnetic biosensors. Can we generalize this remarkable property to non ionizing radiations (NIR) of lower frequencies such as those used by GSM communication devices? We used a Mode Stirred Reverberation Chamber (MSRC, Figure 1) to establish a homogeneous and isotropic electromagnetic field (EMF) that mimics those present in the environment. When plants are subjected to a short, low amplitude, high frequency NIR electromagnetic field (900 MHz, 5 V/m, 10 min), the accumulation of stress-related gene transcripts, such as LebZIP1 (Stankovic et al., 2000) is rapidly, strongly and transiently enhanced (Roux et al., 2005, 2006; Vian et al., 2006, Figure 2). This phenomenon is used to report the reception of the signal. This result imply that coupling mechanisms occurs between EMF and the plant, this last one acting as a living antenna. We propose that the plant conducting system (namely xylem and phloem), that is filled with a diluted electrolytic solution, constitute the central part of the biological reception device. This aspect will be detailed in the final paper; the understanding of the antenna function will help to conceptualize the plant asan integrative EMF detector.

References :

Roux et al., "Systemic accumulation of bZIP mRNA after Low Amplitude 900 MHz stimulation in plant", 16th International Zurich Symposium on Electromagnetic Compatibility, February 13-18, 2005, CDROM.

Roux et al., Electromagnetic fields (900 MHz) evoke consistent molecular responses in tomato plants. Physiologia Plantarum (2006), in press Stankovic et al.,

Molecular cloning and characterization of a tomato cDNA encoding a systematically wound-inducible bZIP DNA-Binding protein. Planta 212: 60-66.

Vian et al., Microwave irradiation affects gene expression in plants, Plant signaling and behavior (2006), 1(2):67-69.